Radio frequency data communications device

ABSTRACT

A frequency doubler includes a first Gilbert cell, a second Gilbert cell coupled to the first Gilbert cell, a frequency generator configured to apply a first sinusoidal wave to the first Gilbert cell, and a phase shifter applying a sinusoidal wave shifted from the first sinusoidal wave to the second Gilbert cell. A method of doubling frequency without using a feedback loop includes providing a first Gilbert cell, providing a second Gilbert cell coupled to the first Gilbert cell, applying a first sinusoidal wave to the first Gilbert cell, and applying a sinusoidal wave shifted from the first sinusoidal wave to the second Gilbert cell.

CROSS REFERENCE TO RELATED APPLICATION

This is a Continuation of U.S. patent application Ser. No. 09/158,530,now U.S. Pat. No. 6,198,332 filed Sep. 22, 1998, and titled “FrequencyDoubler and Method of Doubling Frequency,” now U.S. Pat. No. 6,198,332,which in turn is a Division of U.S. patent application Ser. No.08/705,043, filed Aug. 29, 1996, now U.S. Pat. No. 6,130,602, whichclaims priority from U.S. Provisional Application 60/017,900, filed May13, 1996, titled “Radio Frequency Data Communication Device”.

COPYRIGHT AUTHORIZATION

A portion of the disclosure of this patent document, including theappended microfiche, contains material which is subject to copyrightprotection. The copyright owner has no objection to the facsimilereproduction by anyone of the patent document or the patent disclosure,as it appears in the Patent and Trademark Office patent file or records,but otherwise reserves all copyright rights whatsoever.

REFERENCE TO MICROFICHE

Appended hereto is a microfiche copy of a software guide entitled“Micron RFID Systems Developer's Guide,” May 2, 1996. This appendix has5 microfiche providing 266 total frames.

TECHNICAL FIELD

This invention relates to radio frequency communication devices. Moreparticularly, the invention relates to radio frequency identificationdevices for inventory control, object monitoring, or for determining theexistence, location or movement of objects.

BACKGROUND OF THE INVENTION

As large numbers of objects are moved in inventory, productmanufacturing, and merchandising operations, there is a continuouschallenge to accurately monitor the location and flow of objects.Additionally, there is a continuing goal to interrogate the location ofobjects in an inexpensive and streamlined manner. Furthermore, there isa need for tag devices suitably configured to mount to a variety ofobjects including goods, items, persons, or animals, or substantiallyany moving or stationary and animate or inanimate object. One way oftracking objects is with an electronic identification system.

One presently available electronic identification system utilizes amagnetic field modulation system to monitor tag devices. An interrogatorcreates a magnetic field that becomes detuned when the tag device ispassed through the magnetic field. In some cases, the tag device may beprovided with a unique identification code in order to distinguishbetween a number of different tags. Typically, the tag devices areentirely passive (have no power supply), which results in a small andportable package. However, this identification system is only capable ofdistinguishing a limited number of tag devices, over a relatively shortrange, limited by the size of a magnetic field used to supply power tothe tags and to communicate with the tags.

Another electronic identification system utilizes an RF transponderdevice affixed to an object to be monitored, in which an interrogatortransmits an interrogation signal to the device. The device receives thesignal, then generates and transmits a responsive signal. Theinterrogation signal and the responsive signal are typicallyradio-frequency (RF) signals produced by an RF transmitter circuit.Since RF signals can be transmitted over greater distances than magneticfields, RF-based transponder devices tend to be more suitable forapplications requiring tracking of a tagged device that may not be inclose proximity to an interrogator. For example, RF-based transponderdevices tend to be more suitable for inventory control or tracking.

BRIEF DESCRIPTION OF THE DRAWINGS

The drawings for this patent are the same as those in U.S. Pat. No.6,130,602, which has been incorporated by reference.

SUMMARY OF THE INVENTION

The invention provides a radio frequency identification devicecomprising an integrated circuit including a receiver, a transmitter,and a microprocessor. The integrated circuit is preferably a monolithicsingle die integrated circuit including the receiver, the transmitter,and the microprocessor. Because the device includes an activetransponder, instead of a transponder which relies on magnetic couplingfor power, the device has a much greater range.

One aspect of the invention provides a radio frequency identificationdevice comprising a monolithic integrated circuit including a receiver,a transmitter which can operate at frequencies above 400 MHz, and amicroprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising a monolithic integrated circuitincluding a receiver, a transmitter which can operate at frequenciesabove 1 GHz, and a microprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising a monolithic integrated circuitincluding a transmitter, a microprocessor, and a receiver which canreceive and interpret signals having frequencies above 400 MHz.

Another aspect of the invention provides a radio frequencyidentification device comprising a monolithic integrated circuitincluding a transmitter, a microprocessor, and a receiver which canreceive and interpret signals having frequencies above 1 Ghz.

Another aspect of the invention provides a radio frequencyidentification device comprising a monolithic integrated circuitincluding a receiver, a microwave transmitter, and a microprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising a monolithic integrated circuitincluding a microwave receiver, a transmitter, and a microprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising a single die including a receiver, atransmitter, and a microprocessor, the die having a size less than90,000 mils². In accordance with a more preferred embodiment of theinvention, the die has a size less than 300×300 mils². In accordancewith a more preferred embodiment of the invention, the die has a sizeless than 37,500 mils². In accordance with a more preferred embodimentof the invention, the die has a size of 209 by 116 mils².

Another aspect of the invention provides a radio frequencyidentification device comprising a single die integrated circuitincluding a receiver, a transmitter, and a microprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising a single die with a single metal layerincluding a receiver, a transmitter, and a microprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising a single die integrated circuitincluding a receiver, a transmitter, and a microprocessor formed using asingle metal layer processing method.

Another aspect of the invention provides a radio frequencyidentification system comprising an integrated circuit including areceiver, and a transmitter; and an antenna coupled to the integratedcircuit, the integrated circuit being responsive to radio frequencysignals of multiple carrier frequencies.

Another aspect of the invention provides a radio frequencyidentification device comprising transponder circuitry formed in amonolithic integrated circuit comprising both transmitting and receivingcircuits of the transponder circuitry; a power supply operablyassociated with the transponder circuitry; and an antenna operablyassociated with the transponder circuitry.

Another aspect of the invention provides a radio frequencyidentification device comprising a monolithic semiconductor integratedcircuit including a receiver and a transmitter; means for applying asupply of power to the integrated circuit device from a battery; andmeans for configuring the integrated circuit to receive and transmitradio frequency signals.

Another aspect of the invention provides a method for producing a radiofrequency identification device, the method comprising the followingsteps: providing a monolithic integrated circuit having a receiver and atransmitter; and providing a package configured to carry the integratedcircuit.

Another aspect of the invention provides a method for adapting a radiofrequency data communication device for use at a desired carrierfrequency for use in a radio frequency identification (RFID) device, themethod comprising the following steps: providing an integrated circuithaving tunable circuitry, the integrated circuit comprising a receiverand a transmitter; configuring the integrated circuit for connectionwith a power supply to enable operation; configuring the integratedcircuit to receive and apply radio frequency signals via an antenna, theantenna and the tunable circuitry cooperating in operation therebetween; and tuning the tunable circuitry and the antenna to realize adesired carrier frequency from a wide range of possible carrierfrequencies. A method for adapting a radio frequency data communicationdevice for use at a desired carrier frequency for use in a radiofrequency identification device, the method comprising the followingsteps: providing an integrated circuit having tunable circuitry, theintegrated circuit comprising a receiver and a transmitter; configuringthe integrated circuit for connection with a power supply to enableoperation; configuring the integrated circuit to receive and apply radiofrequency signals via an antenna, the antenna and the tunable circuitrycooperating in operation there between; and tuning the antenna torealize a desired carrier frequency from a wide range of possiblecarrier frequencies.

Another aspect of the invention provides a radio frequencycommunications device comprising an integrated circuit including atransmitter and a receiver, the integrated circuit including a clockrecovery circuit recovering a clock frequency from a signal received bythe receiver, the clock recovery circuit having a phase lock loopincluding a voltage controlled oscillator, and a loop filter having acapacitor storing a voltage indicative of a frequency at which thevoltage controlled oscillator is oscillating, the integrated circuitusing the voltage stored on the capacitor to generate a clock frequencyfor the transmitter.

Another aspect of the invention provides a method of recovering a clockfrequency from a received radio frequency signal, storing the clockfrequency, and using the clock frequency for radio frequencytransmission by a transmitter, the method comprising: providing a clockrecovery circuit recovering a clock frequency from a signal received bythe receiver, the clock recovery circuit having a phase lock loopincluding a voltage controlled oscillator, and a loop filter having acapacitor; using the clock recovery circuit to recover a clock frequencyfrom a received radio frequency signal; storing on the capacitor avoltage indicative of frequency at which the voltage controlledoscillator is oscillating; using the voltage stored on the capacitor togenerate a clock frequency for use by the transmitter.

Another aspect of the invention provides a method of recovering andstoring a clock frequency from a received radio frequency signal in aradio frequency identification device including a transmitter and areceiver, the method comprising providing a clock recovery circuitrecovering a clock frequency from a signal received by the receiver, theclock recovery circuit having a phase lock loop; using the clockrecovery circuit to recover a clock frequency from a received radiofrequency signal; storing in analog form a value indicative of frequencyat which the voltage controlled oscillator is oscillating; and using theanalog value to generate a clock frequency for use by the transmitter.

Another aspect of the invention provides a radio frequencycommunications device comprising an integrated circuit including atransmitter and a receiver, the transmitter being switchable between abackscatter mode, wherein a carrier for the transmitter is derived froma carrier received from an interrogator spaced apart from the radiofrequency communications device, and an active mode, wherein a carrierfor the transmitter is generated by the integrated circuit itself.

Another aspect of the invention provides a radio frequencycommunications device comprising an integrated circuit including atransmitter and a receiver, the transmitter selectively transmitting asignal using a modulation scheme, the transmitter being switchable fortransmission using different modulation schemes.

Another aspect of the invention provides a method for adaptingmodulation schemes of a radio frequency data communication device in aradio frequency identification device, the method comprising thefollowing steps: providing an integrated circuit having switchingcircuitry, a receiver, a transmitter, and a processor; the integratedcircuit having a plurality of transmitting circuits including a firsttransmitting circuit configured to realize an active transmitter schemeand a second transmitting circuit configured to realize a modulatedbackscatter scheme; configuring the integrated circuit for connectionwith a power supply to enable operation; configuring the integratedcircuit to receive and apply radio frequency signals via an antenna, theantenna and the tunable circuitry cooperating in operation; andswitching the switchable circuitry with respect to the antenna to enableone of the transmitting circuits to realize one of the modulationschemes.

Another aspect of the invention provides a method for adaptingmodulation schemes of a radio frequency data communication device in aradio frequency identification device, the method comprising thefollowing steps: providing an integrated circuit having switchingcircuitry, a receiver, a transmitter, and a processor, the integratedcircuit including a plurality of transmitting circuits, the plurality oftransmitting circuits configured to selectively realize a plurality ofmodulated backscatter schemes; configuring the integrated circuit forconnection with a power supply to enable operation; configuring theintegrated circuit to receive and apply radio frequency signals via anantenna, the antenna and the tunable circuitry cooperating in operation;and switching the transmitting circuits with respect to the antenna toenable one of the transmitting circuits to realize one of the modulationschemes.

Another aspect of the invention provides a radio frequencyidentification device comprising: an integrated circuit including atransmitter and a receiver, the integrated circuit being adapted to beconnected to a battery, and further including a comparator comparing thevoltage of the battery with a predetermined voltage and generating a lowbattery signal if the voltage of the battery is less than thepredetermined voltage.

Another aspect of the invention provides a method for detecting a lowbattery condition in a radio frequency data communication device for usein a radio frequency identification device, the method comprising thefollowing steps: providing an integrated circuit having switchingcircuitry, a receiver, and a transmitter, the integrated circuitincluding a comparator configured to compare the battery voltage with apredetermined voltage and generate a low battery signal if the batteryvoltage is less than the predetermined voltage; configuring theintegrated circuit for connection with the battery to enable operation;configuring the integrated circuit to receive and apply radio frequencysignals via an antenna, the antenna and the tunable circuitrycooperating in operation there between; determining a predeterminedvoltage for the battery; comparing the voltage of the battery with thepredetermined voltage; and generating a low battery signal if thevoltage of the battery is less than the predetermined voltage.

Another aspect of the invention provides a radio frequencycommunications device comprising an integrated circuit including atransmitter and a receiver, the integrated circuit periodically checkingif a radio frequency signal is being received by the receiver, theintegrated circuit further including a timer setting a time period forthe checking, the timer having a frequency lock loop.

Another aspect of the invention provides a radio frequencycommunications device comprising an integrated circuit including atransmitter and a receiver, the integrated circuit being configured toperiodically check if a radio frequency signal is being received by thereceiver, the integrated circuit further including a timer setting atime period for the checking, the timer having a phase lock loop.

Another aspect of the invention provides a method for calibrating aclock in a radio frequency data communication device for use in a radiofrequency identification device, the method comprising the followingsteps: providing an integrated circuit having a receiver and atransmitter, the integrated circuit including a timer having a frequencylock loop configured to set a time period for periodically checking if aradio frequency signal is being received by the receiver; configuringthe integrated circuit for connection with a battery to enableoperation; configuring the integrated circuit to receive and apply radiofrequency signals via an antenna, the antenna and the integrated circuitcooperating in operation therebetween; and periodically checking whethera radio frequency signal is being received by the receiver.

Another aspect of the invention provides a radio frequencyidentification device for receiving and responding to radio frequencycommands from an interrogator transmitting a radio frequency signal, thedevice comprising an integrated circuit including a receiver, atransmitter, and a connection pin, the integrated circuit beingswitchable between a radio frequency receive mode wherein the receiverreceives commands via radio frequency, and a direct receive mode whereincommands are received via the connection pin.

Another aspect of the invention provides a radio frequencyidentification device for receiving and responding to radio frequencycommands from an interrogator transmitting a radio frequency signal, thedevice comprising an integrated circuit including a receiver, atransmitter, and a digital input pin, the integrated circuit beingswitchable between a radio frequency receive mode wherein the receiverreceives commands via radio frequency, and a direct receive mode whereincommands are received digitally via the digital input pin. Anotheraspect of the invention provides a radio frequency identification devicefor receiving and responding to radio frequency commands from aninterrogator transmitting a radio frequency signal, the devicecomprising an integrated circuit including a receiver, a transmitter,and a connection pin, the integrated circuit being switchable between aradio frequency receive mode wherein the receiver receives commands viaradio frequency, and a direct receive mode wherein a modulation signalwithout a carrier is received via the connection pin.

Another aspect of the invention provides a radio frequencyidentification device for receiving and responding to radio frequencycommands from an interrogator transmitting a radio frequency signal, thedevice comprising an integrated circuit including a receiver, atransmitter, and a connection pin, the integrated circuit beingswitchable between a radio frequency transmit mode wherein the receivertransmits responses to the commands via radio frequency, and a directtransmit mode wherein responses are transmitted via the connection pin.

Another aspect of the invention provides a radio frequencyidentification device for receiving and responding to radio frequencycommands from an interrogator transmitting a radio frequency signal, thedevice comprising an integrated circuit including a receiver, atransmitter, and a digital output pin, the integrated circuit beingswitchable between a radio frequency transmit mode wherein the receivertransmits responses to the commands via radio frequency, and a directtransmit mode wherein responses are transmitted digitally via thedigital output pin.

Another aspect of the invention provides a radio frequencyidentification device for receiving and responding to radio frequencycommands from an interrogator transmitting a radio frequency signal, thedevice comprising an integrated circuit including a receiver, atransmitter, and a connection pin, the integrated circuit beingswitchable between a radio frequency transmit mode wherein the receivertransmits responses to the commands via radio frequency, and a directtransmit s mode wherein a modulation signal without a carrier istransmitted via the connection pin.

Another aspect of the invention provides a method comprising thefollowing steps: providing an integrated circuit having a receiver, atransmitter, and a connection pin, the integrated circuit including aswitchable circuit configured to switch between a radio frequencyreceive mode wherein the receiver receives commands via radio frequency,and a direct receive mode wherein commands are received via theconnection pin; configuring the integrated circuit for connection with abattery; configuring the integrated circuit to receive and transmitradio frequency signals via an antenna, the antenna and the integratedcircuit cooperating in operation; and switching to one of the radiofrequency receive mode and the direct receive mode to enable receipt ofradio frequency commands or commands received via the connection pin.Another aspect of the invention provides a method comprising thefollowing steps: providing an integrated circuit having a receiver, atransmitter, and a connection pin, the integrated circuit including aswitchable circuit configured to switch between a radio frequencytransmit mode wherein the transmitter transmits information via radiofrequency, and a direct transmit mode wherein data is transmitted viathe connection pin; configuring the integrated circuit for connectionwith a battery; configuring the integrated circuit to receive andtransmit radio frequency signals via an antenna, the antenna and theintegrated circuit cooperating in operation; and switching to one of theradio frequency transmit mode and the direct transmit mode to enabletransmission of information via radio frequency or via the connectionpin.

Another aspect of the invention provides an integrated circuitcomprising a radio frequency receiver; a unique, non-alterable indiciaidentifying the integrated circuit; and a radio frequency transmitterconfigured to transmit a signal representative of the indicia inresponse to a command received by the receiver.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including areceiver for receiving radio frequency commands from an interrogationdevice, and a transmitter for transmitting a signal identifying thedevice to the interrogator, the transmitter and receiver being formed ona die having a lot number, wafer number, and die number, the integratedcircuit including non-alterable indicia identifying the lot number,wafer number, and die number, the transmitter being configured totransmit the non-alterable indicia in response to a manufacturer'scommand received by the receiver, the transmitted non-alterable indiciabeing different from the identifying signal.

Another aspect of the invention provides a method of tracingmanufacturing process problems by tracing the origin of a defectiveradio frequency identification integrated circuit, the methodcomprising: forming a non-alterable indicia on a die for the integratedcircuit, the indicia representing the wafer lot number, wafer number,and die number on the wafer, the indicia being not readily ascertainableby a user; and causing the integrated circuit to transmit thenon-alterable indicia via radio frequency in response to amanufacturer's command.

Another aspect of the invention provides a method of tracing stolenproperty including a radio frequency identification integrated circuit,the method comprising: forming a non-alterable indicia on a die for theintegrated circuit, the indicia representing the wafer lot number, wafernumber, and die number on the wafer, the indicia being not readilyascertainable by a user; and causing the integrated circuit to transmitthe non-alterable indicia via radio frequency in response to amanufacturer's command.

Another aspect of the invention provides a method of tracingmanufacturing process problems in the manufacture of a radio frequencyintegrated circuit by tracing defect origin, the method comprising thefollowing steps: providing a detectable signature on the integratedcircuit, the signature indicative of one or more of the wafer lotnumber, wafer number, and die number of a die for the integratedcircuit; and enabling the integrated circuit to transmit the signaturevia radio frequency responsive to an inquiry command.

Another aspect of the invention provides a radio frequencyidentification device comprising: an integrated circuit including amicroprocessor, a receiver receiving radio frequency commands from aninterrogation device, and a transmitter transmitting a signalidentifying the device to the interrogator, the integrated circuitswitching between a sleep mode, and a microprocessor on mode, in whichmore power is consumed than in the sleep mode, if the microprocessordetermines that a signal received by the receiver is a radio frequencycommand from an interrogation device.

Another aspect of the invention provides a method for conserving powerduring operation of a radio frequency identification device, the methodcomprising the following steps: providing a receiver, a transmitter,microprocessor, and wake-up circuitry, the wake-up circuitry configuredto selectively supply clock signals to the processor and thus controlpower consumption of the processor; configuring the receiver with anantenna to receive radio frequency signals from an interrogation device;configuring the transmitter to transmit a signal identifying the deviceto the interrogator; selectively enabling powered wake-up of thereceiver to periodically check for presence of radio frequency signals;detecting whether a radio frequency signal is valid; and depending onwhether a radio frequency signal is valid, supplying clock signals tothe processor.

Another aspect of the invention provides a method for conserving powerduring operation of a radio frequency identification device, the methodcomprising the following steps: providing a receiver, a transmitter,microprocessor, and wake-up circuitry, the wake-up circuitry configuredto selectively supply power to the processor; configuring the receiverwith an antenna to receive radio frequency signals from an interrogationdevice; configuring the transmitter to transmit a signal identifying thedevice to the interrogator; selectively enabling powered wake-up of thereceiver to periodically check for presence of radio frequency signals;detecting whether a radio frequency signal is valid; and depending onwhether a radio frequency signal is valid, supplying power signals tothe processor.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including amicroprocessor, a transmitter, and a receiver, the integrated circuitbeing switchable between a sleep mode, and a microprocessor on mode in awhich more power is consumed than in the sleep mode, the integratedcircuit being switched from the sleep mode to the microprocessor on modein response to a direct sequence spread spectrum modulated radiofrequency signal, which has a predetermined number of transitions withina certain period of time, being received by the receiver.

Another aspect of the invention provides a method for conserving powerin a radio frequency identification device, the method comprisingperiodically switching from a sleep mode to a receiver on mode andperforming the following tests to determine whether to further switch toa microprocessor on mode because a valid radio frequency signal ispresent: (a) determining if any radio frequency signal is present and,if so, proceeding to step (b); and, if not, returning to the sleep mode;and (b) determining if the radio frequency signal has a predeterminednumber of transitions per a predetermined time period of time and, ifso, switching to the microprocessor on mode; and, if not, returning tothe sleep mode.

Another aspect of the invention provides a radio frequencyidentification device switchable between a sleep mode and a mode inwhich more power is consumed than in the sleep mode, the radio frequencyidentification device comprising a transponder including a receiver anda transmitter; means for periodically checking whether any radiofrequency signal is being received by the receiver; and means fordetermining if a radio frequency signal has a predetermined number oftransitions within a predetermined period of time.

Another aspect of the invention provides a method for conserving powerin a radio frequency identification device, the method comprisingperiodically switching from a sleep mode to a receiver on mode andperforming the following tests to determine whether to further switch toa microprocessor on mode because a valid radio frequency signal ispresent: (a) determining if any radio frequency signal is present and,if so, proceeding to step (b); and, if not, returning to the sleep mode;(b) determining if the radio frequency signal is modulated and has apredetermined number of transitions per a predetermined period of timeand, if so, proceeding to step (c); and, if not, returning to the sleepmode; and (c) determining if the modulated radio frequency signal has apredetermined number of transitions per a predetermined period of timedifferent from the predetermined time of step (b) and, if so, switchingto the microprocessor on mode; and, if not, returning to the sleep mode.

Another aspect of the invention provides a method of forming anintegrated circuit including a Schottky diode, the method comprising:providing a p-type substrate; defining an n-type region relative to thesubstrate; forming an insulator over the n-type region; removing an areaof the insulator for definition of a contact hole, and removing an areaencircling the contact hole; forming n+regions in the n-type regionsencircling the contact hole; depositing a Schottky metal in the contacthole; and annealing the metal to form a silicide interface to the n-typeregion.

Another aspect of the invention provides a method of forming anintegrated circuit including a Schottky diode, the method comprising:providing a substrate; defining a p-type region relative to thesubstrate; forming an insulator over the p-type region; removing an areaof the insulator for definition of a contact hole, and removing an areaencircling the contact hole; forming p+regions in the p-type regionsencircling the contact hole; depositing a Schottky metal in the contacthole; and annealing the Schottky metal to form a silicide interface tothe p-type region.

Another aspect of the invention provides a method of forming anintegrated circuit including a Schottky diode, the method comprising:providing a p-type substrate; defining an n-well region relative to thesubstrate; forming a BPSG insulator over the n-well region; etching awayan area of the BPSG for definition of a contact hole, and etching anarea encircling the contact hole; forming n+regions in the n-wellregions encircling the contact hole; depositing titanium in the contacthole; and annealing the titanium to form a silicide interface to then-well region.

Another aspect of the invention provides a method of forming anintegrated circuit including a Schottky diode, the method comprising:providing an n-type substrate; defining a p-well region relative to thesubstrate; forming a BPSG insulator over the p-well region; etching awayan area of the BPSG for definition of a contact hole, and etching anarea encircling the contact hole;

forming p+regions in the p-well regions encircling the contact hole;depositing titanium in the contact hole; and annealing the titanium toform a silicide interface to the p-well region.

Another aspect of the invention provides a radio frequencycommunications system comprising an antenna; an integrated circuitincluding a receiver having a Schottky diode detector including aSchottky diode coupled to the antenna; and a current source connected todrive current through the antenna and the Schottky diode.

Another aspect of the invention provides an integrated circuit for radiofrequency communications comprising an inductorless radio frequencydetector.

Another aspect of the invention provides a system comprising an antenna;a transponder including a receiver having a Schottky diode detectorincluding a Schottky diode having a first terminal coupled to theantenna and having a second terminal; and means for driving currentthrough both the antenna and the Schottky diode in a direction from thefirst terminal to the second terminal.

Another aspect of the invention provides a system comprising an antenna;

a transponder including a receiver having a Schottky diode detectorincluding a Schottky diode having a first terminal coupled to theantenna and having a second terminal; and means for driving currentthrough both the antenna and the Schottky diode in a direction from thesecond terminal to the first terminal. Another aspect of the inventionprovides a system comprising an antenna;

a transponder including a receiver having a Schottky diode detectorincluding a Schottky diode having an anode coupled to the antenna andhaving a cathode; and means for driving current through both the antennaand the Schottky diode in a direction from the anode to the cathode.

Another aspect of the invention provides a radio frequencycommunications system comprising: an antenna; an integrated circuitincluding a receiver having a Schottky diode detector including aSchottky diode having an anode coupled to the antenna and having acathode, the Schottky diode detector further including a capacitorconnected between the cathode and ground, and including a capacitorhaving a first contact connected to the cathode and having a secondcontact defining an output of the Schottky diode detector; a currentsource connected to the cathode to drive current through the antenna andthe Schottky diode in a direction from the anode to the cathode.

Another aspect of the invention provides a radio frequencycommunications system comprising an antenna; an integrated circuitincluding a receiver having a Schottky diode detector including aSchottky diode having a cathode coupled to the antenna and having ananode, the Schottky diode detector further including a capacitorconnected between the anode and ground, and including a capacitor havinga first contact connected to the anode and having a second contactdefining an output of the Schottky diode detector; and a current sourceconnected to the anode to drive current through the antenna and theSchottky diode in a direction from the anode to the cathode.

Another aspect of the invention provides a system comprising an antenna;a transponder including a receiver having a Schottky diode detectorincluding a Schottky diode having a cathode coupled to the antenna andhaving an anode; and means for driving current through both the antennaand the Schottky diode in a direction from the anode to the cathode.

Another aspect of the invention provides a method for realizing animproved radio frequency detector for use in a radio frequencyidentification device, the method comprising the following steps:providing an integrated circuit and an antenna, the integrated circuithaving a receiver and a transmitter, the integrated circuit furtherhaving a Schottky diode and a current source, with the Schottky diode inoperation being coupled to the antenna and the current source, theSchottky diode and antenna cooperating there between to form aninductorless radio frequency detector; applying a supply of power to theintegrated circuit device from a battery; and applying a desired currentacross the Schottky diode to impart a desired impedance there acrossrelative to the impedance of the antenna.

Another aspect of the invention provides a frequency lock loopcomprising a current controlled oscillator including a plurality ofselectively engageable current mirrors, the frequency of oscillation ofthe frequency lock loop varying in response to selection of the currentmirrors, the current mirrors including transistors operating in asubthreshold mode.

Another aspect of the invention provides an integrated circuitcomprising a receiver, a transmitter, and a frequency lock loopincluding a current source having a thermal voltage generator, a currentcontrolled oscillator having a plurality of selectively engageablecurrent mirrors multiplying up the current of the current source, thefrequency of oscillation of the frequency lock loop varying in responseto selection of the current mirrors, the current mirrors includingtransistors operating in a subthreshold mode.

Another aspect of the invention provides a timing oscillator thatconsumes less than one milliAmp.

Another aspect of the invention provides a method of constructing afrequency lock loop including a current controlled oscillator having aplurality of selectively engageable current mirrors, the frequency ofoscillation of the frequency lock loop varying in response to selectionof the current mirrors, the method comprising selecting current mirrorsto vary frequency of operation, and operating transistors in the currentmirrors in subthreshold mode.

Another aspect of the invention provides a method of operating anintegrated circuit including a receiver, a transmitter, and a frequencylock loop including a current source having a thermal voltage generator,a current controlled oscillator having a plurality of selectivelyengageable current mirrors multiplying up the current of the currentsource, the frequency of oscillation of the frequency lock loop varyingin response to selection of the current mirrors, the method comprisingengaging selected current mirrors and operating transistors in thecurrent mirrors in a subthreshold mode.

Another aspect of the invention provides an amplifier powered by aselectively engageable voltage source, the amplifier comprising firstand second electrodes for receiving an input signal to be amplified, theinput electrodes being adapted to be respectively connected to couplingcapacitors; a differential amplifier having inputs respectivelyconnected to the first and second electrodes, and having an output;selectively engageable resistances between the voltage source andrespective inputs of the differential amplifier and defining, with thecoupling capacitors, the high pass characteristics of the circuit; andsecond selectively engageable resistances between the voltage source andrespective inputs of the differential amplifier, the second resistancesrespectively having smaller values that the first mentioned resistances,the second resistances being engaged then disengaged in response to thevoltage source being engaged.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including amicroprocessor, a receiver receiving radio frequency commands from aninterrogation device, and a transmitter transmitting a signalidentifying the device to the interrogator, the integrated circuitswitching between a sleep mode, and a microprocessor on mode, in whichmore power is consumed than in the sleep mode, if the microprocessordetermines that a signal received by the receiver is a radio frequencycommand from an interrogation device, the integrated circuit furtherincluding an amplifier powered by a selectively engageable voltagesource engaged in the microprocessor on mode but not in the sleep mode,the amplifier including first and second electrodes for receiving aninput signal to be amplified, the input electrodes being adapted to berespectively connected to coupling capacitors, a differential amplifierhaving inputs respectively connected to the first and second electrodes,and having an output, selectively engageable resistances between thevoltage source and respective inputs of the differential amplifier,second selectively engageable resistances between the voltage source andrespective inputs of the differential amplifier, the second resistancesrespectively having smaller values that the first mentioned resistances,the second resistances being engaged then disengaged in response to theintegrated circuit switching from the sleep mode to the microprocessoron mode. Another aspect of the invention provides a method of speedingpower up of an amplifier stage powered by a selectively voltage sourceand including first and second electrodes for receiving an input signalto be amplified, the input electrodes being adapted to be respectivelyconnected to coupling capacitors; a differential amplifier having inputsrespectively connected to the first and second electrodes, and having anoutput; and selectively engageable resistances between the voltagesource and respective inputs of the differential amplifier, the methodcomprising: shorting around the selectively engageable resistances for apredetermined amount of time in response to the voltage source beingengaged.

Another aspect of the invention provides a radio frequencycommunications system comprising an antenna; an integrated circuitincluding a receiver having a Schottky diode detector including aSchottky diode having an anode coupled to the antenna and having acathode, the Schottky diode detector further including a capacitorconnected between the cathode and ground, and including a capacitorhaving a first contact connected to the cathode and having a secondcontact defining an output of the Schottky diode detector, theintegrated circuit further including a clock recovery circuit recoveringa clock from rising edges only of a signal at the output of the Schottkydiode detector; and a current source connected to drive current throughthe antenna and the Schottky diode in a direction from the anode to thecathode.

Another aspect of the invention provides a radio frequencycommunications system comprising an antenna; an integrated circuitincluding a receiver having a Schottky diode detector including aSchottky diode having a cathode coupled to the antenna and having ananode, the Schottky diode detector further including a capacitorconnected between the anode and ground, and including a capacitor havinga first contact connected to the anode and having a second contactdefining an output of the Schottky diode detector, the integratedcircuit further including a clock recovery circuit recovering a clockfrom falling edges only of a signal at the output of the Schottky diodedetector; and a current source connected to drive current through theantenna and the Schottky diode in a direction from the anode to thecathode.

Another aspect of the invention provides a method of recovering a clockin a radio frequency communications system, the method comprising:providing an antenna; providing a receiver having a Schottky diodedetector including a Schottky diode having an anode coupled to theantenna and having a cathode, the Schottky diode detector furtherincluding a capacitor connected between the cathode and ground, andincluding a capacitor having a first contact connected to the cathodeand having a second contact defining an output of the Schottky diodedetector; driving current through the antenna and the Schottky diode ina direction from the anode to the cathode; and recovering a clock fromrising edges only of a signal at the output of the Schottky diodedetector.

Another aspect of the invention provides a method of recovering a clockin a radio frequency communications system, the method comprising:providing an antenna; providing a receiver having a Schottky diodedetector including a Schottky diode having a cathode coupled to theantenna and having an anode, the Schottky diode detector furtherincluding a capacitor connected between the anode and ground, andincluding a capacitor having a first contact connected to the anode andhaving a second contact defining an output of the Schottky diodedetector; driving current through the antenna and the Schottky diode ina direction from the anode to the cathode; and recovering a clock fromfalling edges only of a signal at the output of the Schottky diodedetector.

Another aspect of the invention provides a stage for a voltagecontrolled oscillator, the stage comprising a first transistor having acontrol electrode defining a first input, and having first and secondpower electrodes, the first power electrode defining a first node; asecond transistor having a control electrode defining a second input,and having first and second power electrodes, the first power electrodeof the second transistor defining a second node; a current sourceconnected to the second power electrodes of the first and secondtransistors and directing current away from the second power electrodesof the first and second transistors; and means defining a variableresistance connecting the first and second nodes to a supply voltage.Another aspect of the invention provides a stage for a voltagecontrolled oscillator, the stage comprising a first p-channel transistorhaving a gate defining a control node, having a source adapted to beconnected to a supply voltage, and having a drain; a second p-channeltransistor having a gate connected to the control node, having a sourceconnected to the supply voltage, and having a drain; a first n-channeltransistor having a gate defining a first input, having a drainconnected to the drain of the first p-channel transistor and defining afirst node, and having a source; a second n-channel transistor having agate defining a second input, having a drain connected to the drain ofthe second p-channel transistor and defining a second node, and having asource; a current source connected to the sources of the first andsecond n-channel transistors directing current from the sources of thefirst and second n-channel transistors; a first resistor connectedbetween the supply voltage and the drain of the first n-type transistor;a second resistor connected between the supply voltage and drain of thesecond n-type transistor; a first source follower having an inputconnected to the first node and having an output defining a first outputof the stage; and a second source follower having an input connected tothe second node and having an output defining a second output of thestage.

Another aspect of the invention provides a transmitter including a ringoscillator having a chain of stages, each stage comprising a firstp-channel transistor having a gate defining a control node, having asource adapted to be connected to a supply voltage, and having a drain;a second p-channel transistor having a gate connected to the controlnode, having a source connected to the supply voltage, and having adrain; a first n-channel transistor having a gate defining a firstinput, having a drain connected to the drain of the first p-channeltransistor and defining a first node, and having a source; a secondn-channel transistor having a gate defining a second input, having adrain connected to the drain of the second p-channel transistor anddefining a second node, and having a source; a current source connectedto the sources of the first and second n-channel transistors directingcurrent from the sources of the first and second n-channel transistors;a first resistor connected between the supply voltage and the drain ofthe first n-type transistor; a second resistor connected between thesupply voltage and drain of the second n-type transistor; a first sourcefollower having an input connected to the first node and having anoutput defining a first output of the stage; and a second sourcefollower having an input connected to the second node and having anoutput defining a second output of the stage.

Another aspect of the invention provides a method of varying frequencyin a stage of a voltage controlled oscillator having two inputtransistors having gates defining input nodes and having drain to sourcepaths adapted to be connected between a supply voltage and a currentsource, the method comprising providing an impedance between the inputtransistors and the supply voltage, and varying the impedance.

Another aspect of the invention provides a frequency doubler comprisinga first Gilbert cell; a second Gilbert cell coupled to the first Gilbertcell; a frequency generator configured to apply a first sinusoidal waveto the first Gilbert cell; and a phase shifter applying a sinusoidalwave shifted from the first sinusoidal wave to the second Gilbert cell.

Another aspect of the invention provides a frequency doubler comprisinga first Gilbert cell including a first pair of transistors havingsources that are connected together, a second pair of transistors havingsources that are connected together, a first one of the transistors ofthe first pair having a gate defining a first input node and a first oneof the transistors of the second pair having a gate connected to thefirst input node, a second one of the transistors of the first pairhaving a gate defining a second input node and a second one of thetransistors of the second pair having a gate connected to the secondinput node, the first transistor of the first pair having a drain, andthe second transistor of the second pair having a drain connected to thedrain of the first transistor of the first pair, the second transistorof the first pair having a drain, and the first transistor of the secondpair having a drain connected to the drain of the second transistor ofthe first pair, a third pair including first and second transistorshaving sources coupled together, the first transistor of the third pairhaving a drain connected to the source of the second transistor of thefirst pair, the second transistor of the third pair having a drainconnected to the source of the second transistor of the second pair, anda current source connected to the sources of the third pair and forwardbiasing the third pair, the second transistor of the third pair having agate defining a third input node, and the first transistor of the thirdpair having a gate defining a fourth input node; and a second Gilbertcell including a first pair of transistors having sources that areconnected together, a second pair of transistors having sources that areconnected together, a first one of the transistors of the first pair ofthe second cell having a gate defining a first input node and a firstone of the transistors of the second pair of the second cell having agate connected to the first input node of the second cell, a second oneof the transistors of the first pair of the second cell having a gatedefining a second input node of the second cell and a second one of thetransistors of the second pair of the second cell having a gateconnected to the second input node of the second cell, the firsttransistor of the first pair of the second cell having a drain, and thesecond transistor of the second pair of the second cell having a drainconnected to the drain of the first transistor of the first pair of thesecond cell, the second transistor of the first pair of the second cellhaving a drain, and the first transistor of the second pair of thesecond cell having a drain connected to the drain of the secondtransistor of the first pair of the second cell, a third pair includingfirst and second transistors having sources coupled together, the firsttransistor of the third pair of the second cell having a drain connectedto the source of the second transistor of the first pair of the secondcell, the second transistor of the third pair of the second cell havinga drain connected to the source of the second transistor of the secondpair of the second cell, and a current source connected to the sourcesof the third pair of the second cell and forward biasing the third pairof the second cell, the second transistor of the third pair of thesecond cell having a gate defining a third input node of the secondcell, and the first transistor of the third pair of the second cellhaving a gate defining a fourth input node of the second cell; the drainof the second transistor of the first pair of the second cell beingconnected to the drain of the second transistor of the first pair of thefirst cell, the drain of the second transistor of the second pair of thesecond cell being connected to the drain of the second transistor of thesecond pair of the second cell, the first input node of the second cellbeing connected to the fourth input node of the first cell, the thirdinput node of the second cell being connected to the second input nodeof the first cell, and the fourth input node of the second cell beingconnected to the first input node of the first cell.

Another aspect of the invention provides a method of doubling frequencywithout using a feedback loop, the method comprising: providing a firstGilbert cell; providing a second Gilbert cell coupled to the firstGilbert cell; applying a first sinusoidal wave to the first Gilbertcell; and applying a sinusoidal wave shifted from the first sinusoidalwave to the second Gilbert cell. Another aspect of the inventionprovides a pseudo random number generator comprising a linear feedbackshift register switchably operable in a first mode, and in a second modewherein the shift register consumes more power than in the first mode.

Another aspect of the invention provides a method of generating a pseudorandom number, the method comprising providing a linear feedback shiftregister; providing an oscillator which generates clock signals used bythe linear feedback shift register for shifting; and providing a firstpower level to the oscillator when a pseudo random number is required,and providing a second power level, lower than the first power level, tothe oscillator at other times. Another aspect of the invention providesa method of generating a pseudo random number, the method comprising:providing a linear feedback shift register; providing an oscillatorwhich generates clock signals used by the linear feedback shift registerfor shifting; and operating the oscillator at a first frequency inresponse to a request for a pseudo random number, and operating theoscillator at a second frequency lower than the first frequency afterthe pseudo random number is generated.

Another aspect of the invention provides a system comprising amicroprocessor operating at a frequency; a linear feedback shiftregister operable in a low power mode, wherein the shift registeroperates at a frequency below the frequency of the microprocessor, and ahigh power mode wherein the shift register consumes more power than inthe low power mode, operates at the frequency of the microprocessor, andshifts data into the microprocessor.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including areceiver, a transmitter, a thermal voltage generator, a microprocessoroperating at a frequency, a linear feedback shift register operable in alow power mode, wherein the shift register operates at a frequency belowthe frequency of the microprocessor, and a high power mode wherein theshift register consumes more power than in the low power mode, operatesat the frequency of the microprocessor, and shifts data into themicroprocessor, an oscillator supplying clock signals to the shiftregister, and current mirrors supplying current to each stage of theshift register, the current mirrors being referenced to the thermalvoltage generator when the shift register is in the low power mode, and,when the shift register is in the high power mode, connected to a supplyvoltage potential greater than the potential provided by the thermalvoltage generator.

Another aspect of the invention provides a method of generating a pseudorandom number, the method comprising: providing a thermal voltagegenerator, a linear feedback shift register, an oscillator supplyingclock signals to the shift register, and current mirrors supplyingcurrent to each stage of the shift register; referencing the currentmirrors to the thermal voltage generator when no pseudo random number isrequired; and connecting the current mirrors to a supply voltagepotential greater than the potential provided by the thermal voltagegenerator when a pseudo random number is required.

Another aspect of the invention provides an integrated circuitcomprising a receiver and a transmitter sharing a common antenna.

Another aspect of the invention provides a method of using an integratedcircuit including a receiver and a transmitter, the method comprisingconnecting the receiver and transmitter to a common antenna.

Another aspect of the invention provides an integrated circuitcomprising:

a die including a transmitter having an antenna output and a detectorhaving an antenna input; a package housing the die; a first contactconnected to the antenna output and accessible from outside the package;a second contact connected to the antenna input and accessible fromoutside the package; and a short electrically connecting the firstcontact to the second contact outside the package.

Another aspect of the invention provides a method of using an integratedcircuit including a die having a transmitter including an antenna outputand a detector including an antenna output, the integrated circuitfurther including a package housing the die, a first contact connectedto the antenna output and accessible from outside the package, and asecond contact connected to the antenna input and accessible fromoutside the package, the method comprising: electrically shorting thefirst contact to the second contact outside the package.

Another aspect of the invention provides a transceiver comprising anantenna having a first end connected to a bias voltage, and having asecond end; a detector including a Schottky diode having an anodeconnected to the second end of the antenna; and a transmitter having anoutput connected to the second end of the antenna.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including both areceiver and a transmitter; a first antenna connected to the receiver;and a second antenna connected to the transmitter.

Another aspect of the invention provides a transceiver comprising a loopantenna having a first end connected to a bias voltage, and having asecond end; a second antenna; a detector including a Schottky diodehaving an anode connected to the second end of the antenna; and atransmitter having an output connected to the second antenna.

Another aspect of the invention provides a transceiver comprising anantenna having a first end connected to a bias voltage, and having asecond end; a detector including a Schottky diode having an anodeconnected to the second end of the antenna; and an active transmitterhaving an output connected to the second end of the antenna.

Another aspect of the invention provides a transceiver comprising anantenna having a first end, and having a second end; a detectorincluding a Schottky diode having a cathode connected to the second endof the antenna and defining a potential at the second end of theantenna, the first end of the antenna being connected to a potentiallower than the potential of the second end of the antenna; and abackscatter transmitter including a transistor having a first powerelectrode connected to the first end of the antenna, a second powerelectrode connected to the second end of the antenna, and a controlelectrode adapted to have a modulation signal applied thereto.

Another aspect of the invention provides a transceiver in accordancewith claim and further comprising a current source directing current inthe direction from the anode to the cathode.

Another aspect of the invention provides a transceiver comprising a loopantenna having a first end connected to a bias voltage, and having asecond end; a detector including a Schottky diode having an anodeconnected to the second end of the antenna; a backscatter transmitterhaving a first output and having a second output; a capacitor connectedbetween the first output and the first end of the antenna; and acapacitor connected between the second output and the second end of theantenna.

Another aspect of the invention provides a method of configuring atransceiver including a backscatter transmitter having first and secondoutputs, and a detector having a Schottky diode including an anode, themethod comprising: applying a bias voltage to a first end of an antenna;connecting a second end of the antenna to the anode; connecting acapacitor between the first output and the first end of the antenna; andconnecting a capacitor between the second output and the second end ofthe antenna.

Another aspect of the invention provides a method of arranging atransceiver including a backscatter transmitter and a detector having aSchottky diode including a cathode, the method comprising: connecting afirst end of an antenna to a ground potential; connecting a second endof the antenna to the cathode; and connecting a first power electrode ofa transistor to the first end of the antenna; connecting a second powerelectrode connected to the second end of the antenna; and connecting acontrol electrode of the transistor to a modulation signal.

Another aspect of the invention provides a method of determining when aphase lock loop achieves frequency lock relative to a desired frequency,the phase lock loop including a voltage controlled oscillator having acontrol node and oscillating at a frequency responsive to the voltageapplied to the control node, the method comprising: crossing the voltagethat would result in the phase lock loop tracking the desired frequencyin a first direction; crossing the voltage that would result in thephase lock loop tracking the desired frequency in a second directionopposite the first direction; and indicating that frequency lock hasbeen achieved.

Another aspect of the invention provides a method of determining whenfrequency lock occurs relative to a desired frequency, the methodcomprising:

providing a phase lock loop including a voltage controlled oscillatorthat oscillates at a frequency responsive to voltage applied to thevoltage controlled oscillator; applying a voltage to the voltagecontrolled oscillator to produce a frequency of oscillation less thanthe desired frequency; increasing the voltage applied to the voltagecontrolled oscillator using one or more steps of a first size;increasing the voltage applied to the voltage controlled oscillatorusing one or more steps of a second size smaller than the first size;decreasing the voltage applied to the voltage controlled oscillatorusing one or more steps of a third size smaller than the second size;increasing the voltage applied to the voltage controlled oscillatorusing a step of the third size; and indicating that lock has occurred inresponse to the increase of the step of the third size.

Another aspect of the invention provides a method of determining when aphase lock loop achieves frequency lock relative to a desired frequency,the phase lock loop including a voltage controlled oscillator having acontrol node and oscillating at a frequency responsive to the voltageapplied to the control node, the method comprising: increasing thevoltage applied to the control node to a voltage above the voltage thatwould result in the phase lock loop tracking the desired frequency;decreasing the voltage applied to the control node to a voltage belowthe voltage that would result in the phase lock loop tracking thedesired frequency; and increasing the voltage applied to the controlnode and indicating that frequency lock has been achieved.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including amicroprocessor, a transmitter, and a receiver, the integrated circuitperiodically switching between a sleep mode, and a receiver-on mode inwhich more power is consumed than in the sleep mode, and furtherincluding a selectively engageable timer preventing switching from thesleep mode to the receiver-on mode for a predetermined amount of time.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including amicroprocessor, a transmitter, and a receiver, the integrated circuitperiodically switching between a sleep mode, and a receiver-on mode inwhich more power is consumed than in the sleep mode, and furtherincluding means for selectively preventing switching from the sleep modeto the receiver-on mode for a predetermined amount of time.

Another aspect of the invention provides a radio frequencyidentification device comprising an integrated circuit including amicroprocessor, a transmitter, and a receiver, the integrated circuitbeing switchable between a sleep mode, and a mode in which more power isconsumed than in the sleep mode, the integrated circuit being switchedfrom the sleep mode to the mode in which more power is consumed inresponse to a direct sequence spread spectrum modulated radio frequencysignal being received by the receiver which has a predetermined numberof transitions within a certain period of time, the integrated circuitfurther including a selectively engageable timer which preventsswitching from the sleep mode for a period of time regardless of whethera signal is subsequently received by the receiver which has thepredetermined number of transitions within a certain period of time.

Another aspect of the invention provides a method for conserving powerin a radio frequency identification device, the method comprising:periodically switching from a sleep mode to a receiver on mode andperforming tests to determine whether to further switch to amicroprocessor on mode because a valid radio frequency signal ispresent; and selectively disabling the periodic switching from the sleepmode for a predetermined amount of time.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

This disclosure of the invention is submitted in furtherance of theconstitutional purposes of the U.S. Patent Laws “to promote the progressof science and useful arts” (Article 1, Section 8).

Overview of Device

FIG. 1 illustrates a radio frequency data communication device 12embodying the invention. The radio frequency data communication device12 includes an integrated circuit 16, a power source 18 connected to theintegrated circuit 16 to supply power to the integrated circuit 16, andat least one antenna 14 connected to the integrated circuit 16 for radiofrequency transmission and reception by the integrated circuit 16. Forpurposes of this disclosure, including the appended claims, the term“integrated circuit” shall be defined as a combination of interconnectedcircuit elements inseparably associated on or within a continuoussubstrate. For purposes of this disclosure, including the appendedclaims, the term “semiconductive substrate” is defined to mean anyconstruction comprising semiconductive material, including, but notlimited to, bulk semiconductive materials such as a semiconductive wafer(either alone or in assemblies comprising other materials thereon), andsemiconductive material layers (either alone or in assemblies comprisingother materials). For purposes of this disclosure, including theappended claims, the term “substrate” refers to any supportingstructure, including, but not limited to, the semiconductive substratesdescribed above. In the illustrated embodiment, the integrated circuit16 is a monolithic integrated circuit. For purposes of this disclosure,including the appended claims, the term “monolithic integrated circuit”shall be defined as an integrated circuit wherein all circuit componentsare manufactured into or on top of a single chip of silicon. Theintegrated circuit 16 will be described in greater detail below. Thepower source 18 is a battery or other suitable power source.

Housing

The radio frequency data communication device 12 can be included in anyappropriate housing or packaging.

FIG. 2 shows but one example in the form of an employee identificationbadge 10 including the radio frequency data communication device 12, anda card 11 made of plastic or other suitable material. In one embodiment,the radio frequency data communication device 12 is laminated to theback face of the plastic card 11, and the card forms the visible portionof the badge. In another embodiment, the radio frequency datacommunication device 12 is bonded to the back face of the card byembedding it within a thin bond line of epoxy-based material.Alternatively, the radio frequency data communication device 12 isembedded into the plastic card 11. In one embodiment, the front face ofthe badge 10 has visual identification features including an employeephotograph as well as identifying text.

FIG. 3 illustrates but one alternative housing supporting the device 12.More particularly, FIG. 3 shows a miniature housing 20 encasing thedevice 12 to define a tag which can be supported by an object (e.g.,hung from an object, affixed to an object, etc.). The housing 20preferably has the general shape and size, in plan view, of a postagestamp. The embodiment of FIG. 3 also includes a card 21 supporting thedevice 12 in the housing 20. The card 21 is formed of plastic or othersuitable material having a thickness of about 0.040 inches, a width ofabout 1.25 inches, and a height of about 1.25 inches. In one embodiment,the device 12 is bonded to a back face of the card 21 with a thin layerof non-conductive epoxy material that cooperates with the card to definethe housing 20.

Although two particular types of housings have been disclosed, thedevice 12 can be included in any appropriate housing. The device 12 isof a small size that lends itself to applications employing smallhousings, such as cards, miniature tags, etc. Larger housings can alsobe employed. The device 12, housed in any appropriate housing, can besupported from or attached to an object in any desired manner; forexample using double sided tape, glue, lanyards, leash, nails, staples,rivets, or any other fastener. The housing can be sewn on to an object,hung from an object, implanted in an object (hidden), etc.

Antenna

Various configurations are possible for the antenna 14. The integratedcircuit 16 includes a receiver 30 and a transmitter 32 (FIG. 5). In oneembodiment, separate antennas 44 and 46 are provided for receiver andtransmitter of the integrated circuit 16. In another embodiment (FIG.1), a single antenna is shared by the receiver and transmitter sections.In one embodiment, the antenna is defined by conductive epoxy screenedonto a card or housing. In the illustrated embodiment, the antenna isconductively bonded to the integrated circuit via bonding pads.

In an embodiment where a single antenna is employed, that single antennapreferably comprises a folded dipole antenna defining a continuousconductive path, or loop, of microstrip. Alternatively, the antenna canbe constructed as a continuous loop antenna.

Battery

If the power source 18 is a battery, the battery can take any suitableform. Preferably, the battery type will be selected depending on weight,size, and life requirements for a particular application. In oneembodiment, the battery 18 is a thin profile button-type cell forming asmall, thin energy cell more commonly utilized in watches and smallelectronic devices requiring a thin profile. A conventional buttontypecell has a pair of electrodes, an anode formed by one face and a cathodeformed by an opposite face. Exemplary button-type cells are disclosed inseveral pending U.S. patent applications including U.S. patentapplication Ser. No. 08/205,957, “Button-Type Battery Having BendableConstruction and Angled Button-Type Battery,” listing Mark E. Tuttle andPeter M. Blonsky as inventors; now U.S. Pat. No. 5,432,027; U.S. patentapplication Ser. No. 08/321,251, “Button-Type Batteries and Method ofForming Button-Type Batteries,” listing Mark E. Tuttle as inventor; nowU.S. Pat. No. 5,494,495; and U.S. patent application Ser. No.08/348,543, “Method of Forming Button-Type Batteries and a Button-TypeBattery Insulating and Sealing Gasket,” listing Mark E. Tuttle asinventor now U.S. Pat. No. 5,662,718. These patent applications andresulting patents are hereby incorporated by reference. In analternative embodiment, the battery 18 comprises a series connected pairof button type cells. Instead of using a battery, any suitable powersource can be employed.

Overview of Communication Svstem

FIG. 4 illustrates a radio frequency communication system 24 includingthe device 12 and a radio frequency interrogator unit (hereinafter“interrogator”) 26. The device 12 transmits and receives radio frequencycommunications to and from the interrogator 26. Preferably, theinterrogator unit 26 includes an antenna 28, as well as dedicatedtransmitting and receiving circuitry, similar to that implemented on theintegrated circuit 16. The system 24 further includes a host computer 48in communication with the interrogator 26. The host computer 48 acts asa master in a master-slave relationship with the interrogator 26. Thehost computer 48 includes an applications program for controlling theinterrogator 26 and interpreting responses, and a library (“MRL”) ofradio frequency identification device applications or functions. Most ofthe functions communicate with the interrogator 26. These functionseffect radio frequency communication between the interrogator 26 and thedevice 12. These functions are described below in a section titled“Protocol.”

One example of an interrogator implemented in combination with atransponder unit is disclosed in U.S. Pat. No. 4,857,893, herebyincorporated by reference. Generally, the interrogator 26 includes anantenna 28, and transmits an interrogation signal or command 27(“forward link”) via the antenna 28. The device 12 receives the incominginterrogation signal via its antenna 14. Upon receiving the signal 27,the device 12 responds by generating and transmitting a responsivesignal or reply 29 (“return link”). Preferably, the responsive signal 29is encoded with information that uniquely identifies, or labels theparticular device 12 that is transmitting, so as to identify any objector person with which the device 12 is associated.

In the illustrated embodiment in FIG. 4, there is no communicationbetween devices 12. Instead, the devices 12 communicate with theinterrogator 26. FIG. 4 illustrates the device 12 as being in thehousing 20 of FIG. 3. The system 24 would operate in a similar manner ifthe device 12 is provided in a housing such as the housing 10 of FIG. 2,or any other appropriate housing. Multiple devices 12 can be used in thesame field of an interrogator 26 (i.e., within communications range ofan interrogator 26). Similarly, multiple interrogators 26 can be inproximity to one or more of the devices 12.

Various U.S. patent applications, which are incorporated herein byreference, disclose features that are employed in various alternativeembodiments of the invention: 08/092,147, filed Jul. 15, 1993, “Wake UpDevice for a Communications Systems” now abandoned, and continuationapplication 08/424,827, filed Apr. 19, 1995, “Wake Up Device for aCommunications System”; 08/281,384, filed Jul. 27, 1994, “CommunicationSystem now U.S. Pat. No. 5,790,946; Having Transmitter Frequency Controlnow U.S. Pat. No. 5,568,517 ”; 07/990,918, filed Dec. 15, 1992, now U.S.Pat. No. 5,365,551, “Data Communication Transceiver Using IdentificationProtocol”; 07/899,777, filed Jun. 17, 1992, “Radio FrequencyIdentification Device (RFID) and Method of Manufacture, Including anElectrical Operating System and Method,” now abandoned; 07/921,037,filed Jul. 24, 1992, “Anti-Theft Method for Detecting The UnauthorizedOpening of Containers and Baggage,” now abandoned; 07/928,899, filedAug. 12, 1992, “Electrically Powered Postage Stamp or Mailing orShipping Label Operative with Radio Frequency (RF) Communications,” nowabandoned; and 08/032,384, filed on Mar. 17, 1993, “Modulated SpreadSpectrum in RF Identification Systems Method,” now U.S. Pat. No.5,539,775.

The above described system 24 is advantageous over prior art devicesthat utilize magnetic field effect systems because, with the system 24,a greater range can be achieved, and more information can be obtained(instead of just an identification number).

As a result, such a system 24 can be used, for example, to monitor largewarehouse inventories having many unique products needing individualdiscrimination to determine the presence of particular items within alarge lot of tagged products. The system can also be used to counteractterrorism to monitor luggage entering a plane to ensure that each itemof luggage that enters the plane is owned by a passenger who actuallyboards the plane. Such a technique assumes that a terrorist will notboard a plane that he or she is planning to bomb. The system 24 isuseful whenever RF transmission over a large range is desirable, such asfor inventory control. In one embodiment, the sensitivity of the devices12 is adjustable so that only devices within a certain range of theinterrogator 26 will respond. In another embodiment, the power of theinterrogator 26 is adjustable so that only devices within a certainrange of the interrogator 26 will respond.

However, a power conservation problem is posed by such implementationswhere batteries are used to supply power to the integrated circuits 16.If the integrated circuit 16 operates continuously at full power,battery life will be short, and device 12 will have to be frequentlyreplaced. If the battery 18 is permanently sealed in a housing,replacement of the battery will be difficult or impossible. For example,one reason for sealing the battery with the integrated circuit 16 andantenna 14 in a housing is to simplify the design and construction, toreduce the cost of production, and protect the electricalinterconnections between devices. Another reason is protection of thebattery and integrated circuit 16 from moisture and contaminants. Athird reason is to enhance the cosmetic appeal of the device 12 byeliminating the need for an access port or door otherwise necessary toinsert and remove the battery. When the battery is discharged, theentire badge or stamp is then discarded. It is therefore desirable inthis and other applications to incorporate power conservation techniquesinto the integrated circuit 16 in order to extend useful life.

In one embodiment, the devices 12 switch between a “sleep” mode ofoperation, and higher power modes to conserve energy and extend batterylife during periods of time where no interrogation signal 27 is receivedby the device 12. These power conservation techniques are described ingreater detail below.

In one embodiment of the invention, in order to further extend the lifeof the battery 18, the receiver sensitivity of the device 12 is tunedover a range of tuned and detuned states in order to modify the abilityof the device to detect signal 27, and therefore adjust the tendency forthe device to wake up. One way to adjust the receiver sensitivity is byadjusting the sensitivity, or impedance of the antenna. Another way isby controlling the gain of amplifiers included in the receiver. Anotherway is to adjust or switch in different circuit elements in the device12, thereby realizing different circuit configurations. Additionally,the transmitting sensitivity for the device 12 can be adjusted. Forexample, transmitting range can be adjusted by controlling interrogatorcontinuous wave power if the transmitter is operating in backscattermode, and by controlling output power if the transmitter is in activemode.

Overview of Integrated Circuit

FIG. 5 is a high level circuit schematic of the integrated circuit 16utilized in the devices of FIGS. 1-4. In the embodiment shown in FIG. 5,the integrated circuit 16 is a monolithic integrated circuit. Moreparticularly, in the illustrated embodiment, the integrated circuit 16comprises a single die, having a size of 209×116 mils², including thereceiver 30, the transmitter 32, a micro controller or microprocessor34, a wake up timer and logic circuit 36, a clock recovery and datarecovery circuit 38, and a bias voltage and current generator 42.

In one embodiment, a spread spectrum processing circuit 40 is alsoincluded in the integrated circuit 16 and formed relative to the singledie. In this embodiment, signals received by the receiver 30 aremodulated spread spectrum signals. Spread spectrum modulation isdescribed below. In the illustrated embodiment, the modulation schemefor replies sent by the transmitter 32 is selectable. One of theavailable selections for replies sent by the transmitter 32 is modulatedspread spectrum.

Spread Spectrum Modulation

Many modulation techniques minimize required transmission bandwidth.However, the spread spectrum modulation technique employed in theillustrated embodiment requires a transmission bandwidth that is up toseveral orders of magnitude greater than the minimum required signalbandwidth. Although spread spectrum modulation techniques are bandwidthinefficient in single user applications, they are advantageous wherethere are multiple users, as is the case with the instant radiofrequency identification system 24. The spread spectrum modulationtechnique of the illustrated embodiment is advantageous because theinterrogator signal can be distinguished from other signals (e.g.,radar, microwave ovens, etc.) operating at the same frequency. Thespread spectrum signals transmitted by the device 12 and by theinterrogator 26 (FIG. 4) are pseudo random and have noise-likeproperties when compared with the digital command or reply. Thespreading waveform is controlled by a pseudo-noise or pseudo randomnumber (PN) sequence or code (described below). The PN code is a binarysequence that appears random but can be reproduced in a predeterminedmanner by the device 12. More particularly, incoming spread spectrumsignals are demodulated by the device 12 through cross correlation witha version of the pseudo random carrier that is generated by the device12 itself. Cross correlation with the correct PN sequence unspreads thespread spectrum signal and restores the modulated message in the samenarrow band as the original data.

A pseudo-noise or pseudo random sequence (PN sequence) is a binarysequence with an autocorrelation that resembles, over a period, theautocorrelation of a random binary sequence. The autocorrelation of apseudo-noise sequence also roughly resembles the autocorrelation ofband-limited white noise. A pseudo-noise sequence has manycharacteristics that are similar to those of random binary sequences.For example, a pseudo-noise sequence has a nearly equal number of zerosand ones, very low correlation between shifted versions of the sequence,and very low cross correlation between any two sequences. A pseudo-noisesequence is usually generated using sequential logic circuits. Forexample, a pseudo-noise sequence can be generated using a feedback shiftregister.

A feedback shift register comprises consecutive stages of two statememory devices, and feedback logic. Binary sequences are shifted throughthe shift registers in response to clock pulses, and the output of thevarious stages are logically combined and fed back as the input to thefirst stage. The initial contents of the memory stages and the feedbacklogic circuit determine the successive contents of the memory.

The illustrated embodiment employs direct sequence spread spectrummodulation. A direct sequence spread spectrum (DSSS) system spreads thebaseband data by directly multiplying the baseband data pulses with apseudo-noise sequence that is produced by a pseudo-noise generator. Asingle pulse or symbol of the PN waveform is called a “chip.”Synchronized data symbols, which may be information bits or binarychannel code symbols, are added in modulo-2 fashion to the chips beforebeing modulated. The receiver performs demodulation. For example, in oneembodiment the data is phase modulated, and the receiver performscoherent or differentially coherent phase-shift keying (PSK)demodulation. In another embodiment, the data is amplitude modulated.Assuming that code synchronization has been achieved at the receiver,the received signal passes through a wideband filter and is multipliedby a local replica of the PN code sequence. This multiplication yieldsthe unspread signal.

A pseudo-noise sequence is usually an odd number of chips long. In theillustrated embodiment, one bit of data is represented by a thirty-onechip sequence. A zero bit of data is represented by inverting thepseudo-noise sequence.

Spread spectrum techniques are also disclosed in the following patentapplications and patent, which are incorporated herein by reference:U.S. patent application Ser. No. 08/092,147; U.S. patent applicationSer. No. 08/424,827, filed Apr. 19, 1995; and U.S. Pat. No. 5,121,407 toPartyka et al. They are also disclosed, for example, in “Spread SpectrumSystems,” by R. C. Dixon, published by John Wiley and Sons, Inc.

The system disclosed in U.S. patent application Ser. No. 08/092,147includes two receivers, a low power receiver for detecting a wake upsignal from an interrogator, and a high power receiver for receivingcommands from an interrogator. On the other hand, the integrated circuit16 of the illustrated embodiment employs a single receiver for both wakeup and receiving commands from an interrogator. Another difference isthat in the system 12 of the illustrated embodiment the receiver, notthe interrogator, controls wake up. In the system 24 of the illustratedembodiment, the integrated circuit 16 includes a timer that causes thereceiver and support circuitry to be powered on periodically. This isdescribed in greater detail elsewhere.

Backscatter and Frequency Hopping

The interrogator sends out a command that is spread around a certaincenter frequency (e.g, 2.44 GHz). After the interrogator transmits thecommand, and is expecting a response, the interrogator switches to a CWmode (continuous wave mode). In the continuous wave mode, theinterrogator does not transmit any information. Instead, theinterrogator just transmits 2.44 GHz radiation. In other words, thesignal transmitted by the interrogator is not modulated. After thedevice 12 receives the command from the interrogator, the device 12processes the command. If the device 12 is in a backscatter mode italternately reflects or does not reflect the signal from theinterrogator to send its reply. For example, in the illustratedembodiment, two halves of a dipole antenna are either shorted togetheror isolated from each other, as described below in greater detail. Inthe illustrated embodiment, frequency hopping does not occur when theinterrogator transmits a command, but occurs when the interrogator is inthe continuous wave mode. The interrogator, in the continuous wave mode,hops between various frequencies close to the 2.44 GHz frequency. Thesevarious frequencies are sufficiently close to the 2.44 GHz frequencythat backscatter antenna reflection characteristics of the device 12 arenot appreciably altered. Because the interrogator is hopping betweenfrequencies, the interrogator knows what frequency backscatterreflections to expect back from the device 12. By hopping betweenvarious frequencies, the amount of time the interrogator continuouslyuses a single frequency is reduced. This is advantageous in view of FCCregulatory requirements.

In the illustrated embodiment, no attempt is made to frequency hop atthe interrogator to a pseudo-random sequence and then correlate to thatat the receiver. However, in alternative embodiments, such correlationtakes place.

CMOS Process

The integrated circuit 16 is formed according to semiconductor waferprocessing steps, such as CMOS semiconductor wafer processing steps usedto form static random access memories. In the preferred embodiment, theintegrated circuit 16 is a single metal integrated circuit. In otherwords, the integrated circuit 16 is formed using a single metal layerprocessing method. More particularly, only one layer of metal (e.g.,aluminum) is employed. This is advantageous in that it results in alower cost of production.

In this processing method, a p-type wafer is employed. The processingmethod employed provides n-well areas used to define p-channeltransistors; an active area which is used to define p+ and n+ diffusedregions inside the p-type wafer or inside the n-well areas. Next, alayer is provided that helps prevent leakage between adjacent devices.Then, transistor are defined by forming n-type and p-type polysilicon.Then, a contact layer is defined for connecting desired intersections ofpolysilicon with metal (aluminum) that is subsequently formed. Thecontact layer is also used, in some instances, for connecting desiredintersections of the metal that is subsequently formed with active area.Then the metal layer is formed. The contact layer provides a means forconnecting metal with layers below the metal. Then, a passivation stepis performed. Passivation means that the die is covered with aprotective layer and holes are cut around the edge of the die so thatelectrical connection can be made to the bond pads.

In some processing, after the metal layer is formed, an insulating layeris provided, and another layer of aluminum is formed above theinsulating layer. Holes are provided at selected locations tointerconnect the top layer of aluminum with lower layers. An advantageof using multiple layers of metal is that it provides greaterflexibility in how functional blocks are laid out and in how power isbused to various areas. However, multiple metal layers add processingsteps. This results in added cost and complexity.

The process of the preferred embodiment employs only one layer of metal,and is therefore a relatively simple, inexpensive process.

The following U.S. patents, which are incorporated herein by reference,disclose CMOS processing techniques that are employed in variousalternative embodiments of the invention: U.S. Pat. No. 5,489,546 toAhmad et al.; U.S. Pat. No. 5,272,367 to Dennison et al.; and U.S. Pat.No. 5,134,085 to Gilgen et al.

Various other processing methods can be employed in alternativeembodiments.

Transmitter and Receiver

The receiver 30 is a radio frequency receiver included in the integratedcircuit 16, and the transmitter 32 is a radio frequency transmitterincluded in the integrated circuit 16. In one embodiment, the receiver30 includes a Schottky diode detector. Various forms of Schottky diodedetectors are described in a paper titled “Designing Detectors for RF/IDTags,” by Raymond W. Waugh of Hewlett-Packard Company, submitted forpresentation at the RF Expo, San Diego, Feb. 1, 1995, and incorporatedherein by reference.

The receiver 30 of the illustrated embodiment makes use of the rate orfrequency of data included in incoming signals, but does not make use ofthe carrier frequency of the incoming signal. In other words, operationof the receiver 30 is independent of the frequency of the carrier of theincoming signal over a wide range of carrier frequencies.

Therefore, the device 12 can operate over a wide range of carrierfrequencies. For example, the device 12 can operate with carriers of915-5800 MHZ. In a more particular embodiment, the device 12 can operatewith carrier frequencies in the 915, 2450, or 5800 MHZ bands. In theillustrated embodiment, the antennas are half wave antennas, andfrequency selectivity of the device 12 is achieved based on selection ofthe antenna external to the integrated circuit 16. Capacitors employedin the Schottky diode detector are also selected based on the carrierfrequency that will be employed.

In one embodiment, the transmitter 32 is switchable between operating ina modulated backscatter transmitter mode, and operating in an activemode. The transmitter 32 switches between the backscatter mode and theactive mode in response to a radio frequency command, instructing thetransmitter to switch, sent by the interrogator 26 and received by thereceiver 30. In the active mode, a carrier for the transmitter isextracted from a signal received by the receiver 30.

Active transmitters are known in the art. See, for example, U.S. patentapplication Ser. No. 08/281,384. U.S. patent application Ser. No.08/281,384 also discloses how transmit frequency for the transmitter 32is recovered from a message received via radio frequency from theinterrogator 26. The device 12 differs from the device disclosed in U.S.patent application Ser. No. 08/281,384 in that a VCO control voltage isstored as an analog voltage level on a capacitor instead of as a digitalnumber in a register. Further, in the illustrated embodiment, therecovered frequency is also used by the integrated circuit 16 togenerate a DPSK subcarrier for modulated backscatter transmission.

The transmitter is capable of transmitting using different modulationschemes, and the modulation scheme is selectable by the interrogator.More particularly, if it is desired to change the modulation scheme, theinterrogator sends an appropriate command via radio frequency. Thetransmitter can switch between multiple available modulation schemessuch as Binary Phase Shift Keying (BPSK), Direct Sequence SpreadSpectrum, On-Off Keying (OOK), and Modulated Backscatter (MBS).

Wake Up Timer and Logic Circuit

The integrated circuit 16 includes the wake up timer and logic circuit36 for conserving battery power. More particularly, the integratedcircuit 16 normally operates in a sleep mode wherein most circuitry isinactive and there is a very low current draw on the battery 18. Onecircuit that is active during the sleep mode is a timer for waking upthe integrated circuit at predetermined intervals. In the illustratedembodiment, the interval is 16 milliseconds; however, various otherintervals can be selected by radio frequency by sending a message fromthe interrogator 26 to the device 12. For example, in the illustratedembodiment, the interval is selectable as being 0.5, 16, 64 or 256milliseconds.

Assuming the selected interval is 16 milliseconds, after every sleepperiod of 16 milliseconds the wake up timer and logic circuit 36activates the receiver 30, the clock recovery and data recovery circuit38, and all the bias currents and voltages associated with the receiver30. This is a receiver on mode, illustrated by a vertical line markedWAKEUP RX ON in FIG. 27. Such bias currents and voltages are generatedby the bias voltage and current generator 42. The receiver 30 thendetermines if there is a radio frequency signal present.

If there is no radio frequency signal present, the wake up timer andlogic circuit 36 deactivates the receiver 30 and clock recovery and datarecovery circuit 38. The receiver then goes back to sleep in the lowcurrent mode until another 16 milliseconds pass (or whatever sleepperiod is selected).

If there is a radio frequency signal present, the receiver will unspreadthe spread spectrum signal for processing. It is possible that while thereceiver is on, it may detect a radio frequency signal from a sourceother than the interrogator 26. For example, other radio frequencytransmitting devices may be operating in the area. In the illustratedembodiment, the receiver is set to receive microwave frequency signals,so that a small antenna can be used. Therefore, the wake up timer andlogic circuit 36 performs tests to determine if a radio frequency signalreceived on wake up is valid. This is a wake up abort test mode,illustrated by a vertical line marked WAKEUP ABORT TESTS in FIG. 27. Ifthe wake up timer and logic circuit 36 determines that the incomingsignal is not valid, the integrated circuit 16 returns to the sleepmode. The illustrated integrated circuit 16 consumes approximately onemicro amp in the sleep mode, and the battery 18 is expected to last upto 10 years with a current drain of that order, depending on how oftenradio frequency signals are present and on the capacity of the battery.

If a radio frequency signal is detected upon wake up, the wake up timerand logic compares the incoming signal to known characteristics ofexpected spread spectrum encoded data. In the illustrated embodiment, avalid incoming radio frequency signal will be a spread spectrum signalhaving a thirty-one chip code representing a single data bit. Torepresent a digital one (“1”) the thirty-one chip code is sent as is. Torepresent a digital zero (“0”) the thirty-one chip code is inverted. Thewake up timer and logic circuit 36 knows how many transitions there arein a valid thirty-one chip sequence, and knows the time period withinwhich all those transitions are expected (or the frequency of thetransitions). After the incoming radio frequency signal is amplified andconverted to baseband, it is tested against known characteristics of avalid signal.

If the incoming signal does not pass these tests, the integrated circuit16 returns to the sleep mode. If the incoming signal does pass thesetests, then the wake up timer and logic circuit determines whether theclock recovery and data recovery circuit 38 locks on to the clockfrequency contained in the chip rate of the incoming signal within apredetermined time period. If frequency lock is obtained, themicroprocessor is turned on for processing of the received command. thisis a processor on mode illustrated by a vertical line marked “PROCESSORON” in FIG. 27.

If frequency lock is not obtained within the predetermined time, theintegrated circuit 16 returns to the sleep mode.

Other appropriate tests can be performed in embodiments where spreadspectrum is not employed. In these embodiments, knowing how valid datais encoded, the wake up timer and logic still compares the number oftransitions received in a given amount of time with an expected numberof transitions for a valid signal.

In summary, various tests are performed, and the order in which they areperformed is preferably selected to most quickly identify invalidsignals. U.S. patent application Ser. No. 08/424,827, filed Apr. 19,1995 and U.S. patent application Ser. No. 08/092,147, which areincorporated herein by reference, disclose tests that could be employedin various alternative embodiments of the invention.

After the wake up timer and logic circuit 36 determines that a receivedsignal is valid, the integrated circuit 16 then performs clock recovery.To save space and cost, the preferred device 12 does not include acrystal timing element (clock). Instead, all timing for the device 12 isextracted from valid incoming signals received by the receiver 30.

In one embodiment, a valid incoming radio frequency signal is digital,and starts with a preamble, which is followed by a start code (or Barkercode), which is followed by data (e.g., a command). For example, in theillustrated embodiment, the preamble is a long (e.g., eighteenmilliseconds) string of zeros; i.e., the thirty-one chip sequence isinverted, and sent repeatedly for approximately 18 milliseconds. In theillustrated embodiment the data or command after the Barker code isshorter than the preamble, and is approximately 4 milliseconds long.

Clock Recovery and Data Recovery Circuit

The clock for the entire integrated circuit 16 is extracted from theincoming message itself. In one embodiment, the transmitter 32 isselectable as being operable in an active transmission mode, or abackscatter mode. If the transmitter 32 is operating in an active mode,the extracted clock is multiplied up to the carrier frequency of thetransmitter 32. For example, in one embodiment, the transmitter carrierfrequency is 2.44 GHz. The choice of chip rate is a function of thecarrier frequency and the carrier frequency has to be divisible by apower of two to give the chip rate on the input.

If the transmitter 32 is operating in a backscatter mode, the clock thathas been recovered from the incoming signal received by the receiver 30is divided to make it slower and is then used for frequency shift key orphase shift key modulated backscatter.

In summary, a clock is recovered from the incoming message, and used fortiming for the micro controller 34 and all the other clock circuitry onthe chip, and also for deriving the transmitter carrier or thesubcarrier, depending on whether the transmitter is operating in activemode or backscatter mode.

Note that there are disadvantages to generating a transmit frequency inthis fashion. In an alternative embodiment (not shown), a crystal isemployed to generate a clock. A crystal provides a more stable, reliableclock to generate the transmit frequency, but also increases cost andsize of the device 12.

In addition to recovering a ciock, the clock recovery and data recoverycircuit 38 also performs data recovery on valid incoming signals. Thevalid spread spectrum incoming signal is passed through the spreadspectrum processing circuit 40, and the spread spectrum processingcircuit 40 extracts the actual ones and zeros of data from the incomingsignal. More particularly, the spread spectrum processing circuit 40takes the chips from the spread spectrum signal, and reduces eachthirty-one chip section down to a bit of one or zero, which is passed tothe micro controller 34.

Micro Controller

The micro controller 34 includes a serial processor, or I/O facilitythat received the bits from the spread spectrum processing circuit 40.The micro controller 34 performs further error correction. Moreparticularly, a modified hamming code is employed, where each eight bitsof data is accompanied by five check bits used by the micro controller34 for error correction. The micro controller 34 further includes amemory, and after performing the data correction, the micro controller34 stores bytes of the data bits in memory. These bytes contain acommand sent by the interrogator 26. The micro controller 34 responds tothe command.

For example, the interrogator 26 may send a command requesting that anydevice 12 in the field respond with the device's identification number.Status information is also returned to the interrogator 26 from thedevice 12 when the device 12 responds.

Unalterable Identification

In one embodiment, the integrated circuit 16 includes unalterableindicia (a signature), different from the device's identification numberdiscussed above. The unalterable indicia is burned into programmableread only memory or formed using a laser operating on fusible links. Theunalterable indicia is indicative of the history of the particular dieused to manufacture the integrated circuit 16. For example, in theillustrated embodiment, the unalterable indicia includes a lot number,wafer number, and die number of the die used to manufacture theintegrated circuit 16. This information is transmitted by thetransmitter in response to a manufacturer's command received by thereceiver. In one embodiment, the manufacturer's command is a controlledaccess, or secret command that is not readily ascertainable by thepublic or purchaser/user of the device. This unalterable indicia can beused to trace manufacturing problems in defective devices 12, or tolocate stolen products carrying a device 12.

Arbitration

If the interrogator 26 sends out a command requesting that all devices12 within range identify themselves, and gets a large number ofsimultaneous replies, the interrogator 26 may not able to interpret anyof these replies. Further, there may be multiple interrogators in anarea trying to interrogate the same device 12.

Therefore, arbitration schemes are provided. With the more commonscenario of multiple devices 12 trying to respond to an interrogator,the interrogator 26 sends a command causing each device 12 of apotentially large number of responding devices 12 to select a randomnumber from a known range and use it as that device's arbitrationnumber. By transmitting requests for identification to various subsetsof the full range of arbitration numbers, and checking for an error-freeresponse, the interrogator 26 determines the arbitration number of everyresponder station capable of communicating at the same time. Therefore,the interrogator 26 is able to conduct subsequent uninterruptedcommunication with devices 12, one at a time, by addressing only onedevice 12.

If the interrogator 26 has prior knowledge of the identification numberof a device 12 which the interrogator 26 is looking for, it can specifythat a response is requested only from the device 12 with thatidentification number.

Arbitration schemes are discussed below, in greater detail, inconnection with protocols.

U.S. Pat. No. 5,365,551 to Snodgrass et al., which is incorporated byreference, discloses arbitration schemes that could be employed invarious alternative embodiments of the invention.

Reply

After the micro controller processes a command from the interrogator 26,the micro controller formats the reply as specified in the protocol andthe formatted reply leaves the micro controller via a serial data portof the micro controller. If desired, the formatted reply is spreadspectrum encoded by the spread spectrum processing circuit 40. The replyis then modulated by the transmitter 32. The transmitter 32 is capableof transmitting using different modulation schemes, and the modulationscheme is selectable by the interrogator 26. More particularly, if it isdesired to change the modulation scheme, the interrogator 26 sends anappropriate command via radio frequency.

The transmitted replies have a format similar to the format of incomingmessages. More particularly, a reply starts with a preamble (e.g., allzeros in active mode, or alternating double zeros and double ones inbackscatter mode), followed by a Barker or start code which is thirteenbits long, followed by actual data.

No stop bits are included in the incoming message or reply, in thepreferred embodiment. Instead, part of the incoming message describeshow many bytes are included, so the integrated circuit 16 knows how muchinformation is included. Similarly, part of the outgoing reply describeshow many bytes are included, so the interrogator 12 knows how muchinformation is included. The incoming message and outgoing replypreferably also include a check sum or redundancy code so that theintegrated circuit 16 or the interrogator 12 can confirm receipt of theentire message or reply.

After the reply is sent, the integrated circuit 16 returns to the sleepmode, and the wake up timer and logic circuit 36 starts timing again forthe next wake up (e.g., in 16 milliseconds, or whatever period isselected).

Detailed Circuit Schematics

FIG. 6 is a graph illustrating how FIGS. 6AA-EK are to be assembled.

FIGS. 6AA-EK include circuitry partitioned in blocks in a manner that issomewhat different from the way the blocks are partitioned in FIG. 5. Insome ways FIGS. 6AA-EK shows less detail than in FIG. 5, and in someways they show more detail.

The integrated circuit 16 is shown as including an analog processor“anlgproc,” an RF processor “rfproc,” a PN (pseudo random number)processor “pnproc,” a data processor “dataproc,” and return linkconfiguration logic “rlconfig.”

The data processor “dataproc” shown in FIGS. 6AA-EK is the microcontroller or microprocessor 34 of FIG. 5. The data processor “dataproc”is shown in greater detail in FIG. 7. In the illustrated embodiment, thedata processor “dataproc” is an eight bit processor, and includes a ROM“rom,” a RAM “ram,” a serial I/O block “sio,” an eight bit ALU(arithmetic logic unit) “alu,” an instruction decoder programmable logicarray “insdec,” and address decoder “adrdec,” a clock generator “clk,” aconditional qualifier decoder “cqualdec,” a databus latch/prechargecircuit “dblatch,” a timed lockout divider “tld,” a data interleaver(which interleaves two thirteen bit words) “dil,” a convolutionalencoder and preamble generator “conv,” a digital port output controller“doutport,” a shift register input data multiplexer “shdel” and a seriesof registers. In the illustrated embodiment, the registers include atimed lockout register “tloreg,” a plurality of status registers “sreg,”a plurality of read/write control registers “oreg,” and an instructionregister “insreg.”

The registers are used to drive control lines to various differentcircuits to allow the data processor to have control over thosecircuits. The “sio” block (described below) is the data path for datareceived and for the data to be transmitted.

FIG. 6.01 is a layout diagram illustrating the physical layout ofvarious components on an integrated circuit die, in accordance with oneembodiment of the invention. The physical locations and sizes ofcomponents relative to other components are shown. Boundaries betweenvarious blocks may be approximate in the sense that portions of certainblocks may extend into other blocks. The layout diagram illustrates thatseparate analog and digital ground returns are provided. In theillustrated embodiment, the ground return for the receiver andtransmitter is spaced apart from the receiver and transmitter. However,in an alternative embodiment, locating the ground return for thereceiver and transmitter proximate the receiver and transmitter mayprovide improved results. In the preferred embodiment, the transmitterand receiver circuitry is physically located on the die close to anedge, proximate to the bond pads. More particularly, the microwaveoutputs of the transmitter 32 are arranged on the die so as to be nextto (in close physical proximity to) the appropriate bond pads. Alsoshown in FIG. 6.01 are small squares adjacent the receiver and activetransmitter pads, respectively. These are ground pads for microwaveprobing, in the exemplary embodiment. In an alternative embodiment,these microwave probing ground pads can be employed as functional groundpads instead of using the illustrated common analog ground pad.

FIG. 7.01AA-BB provide a circuit drawing of a processor clock generator“clk.” The processor clock generator provides clock circuitry thatgenerates all the various clocks that are used by the processor.

FIGS. 7.0101AA-BB provide a circuit drawing of a processor clockcontroller “clkctl.” The clock controller “clkctl” determines when theclocks are running. As described elsewhere, the processor is not alwayson. The clock controller uses enabling signals from wake up so that itknows when to turn on. Thus, some of the inputs to the clock controllerare power wake up, receive wake up, timer wake up. The clock controlleralso synchronizes shut down of the clocks when the processor hascompleted its task.

FIGS. 7.0102AE-DJ provide a circuit drawing of a processor phasegenerator “clkph.” The processor phase generator “clkph” generatesmaster clocks—phase one “PH1” and phase two “PH2”—which arenon-overlapping clocks.

FIGS. 7.0103AA-BD provide a circuit drawing of a clock state generator“clkst.” The clock state generator “clkst” generates some derivativeclocks. Processor instruction cycles are divided. There are cycles andthere are states. Within each cycle, which is a certain time period,there are four states—S1 through S4. The states are all non-overlapping,and each state has a high time that is one quarter of the cycle time. Asa processor instruction executes, the instruction is taken from the rom“rom,” and loaded into the instruction register. The instruction can be,for example, a 1, 2 or 3 cycle instruction, depending on how complex thefunction is that is performed by that instruction. These are microinstructions for running the processor on chip. They should not beconfused with the commands that are sent by radio frequency, which are amuch higher level commands. The commands sent by radio frequency requiremany of these micro instructions for the processor to carry them out.During clock cycle one, line C1 in FIGS. 7.0103AA-BD is high, duringclock cycle two, line C2 is high, etc. Within each one of those clockcycles, state one is high for a certain time period and then goes low,and state two goes high for a certain time period then goes low, and soon up through state four. Within each of these states, there is onephase one high time, and one phase two high time.

FIGS. 7.02AA-BF provide a circuit drawing of an address decoder“adrdec.” In executing instructions, the processor has the need to movebytes of data between registers and ram “ram” and possibly to the serialIO controller “sio.” The address decoder “adrdec” generates enable linesto those various different blocks (the registers, ram, and sio, asappropriate) when their address appears on the address bus. The primaryinput to the address decoder is the address bus. This decoder decideswhich circuit block is being addressed and issues an enable for either awrite or a read, whichever is appropriate, to that particular block.

FIGS. 7.03AA-EH provide a circuit drawing of random access memory “ram.”The ram has 512 bytes of storage available. 256 of those bytes areavailable to the user of the device 12, and the other 256 bites are usedto do calculations required by the processor. Most of the drawing istaken up by blocks of RAM arrays “ram8×4.”

FIGS. 7.0301AA-BB provide a circuit drawing of a ram control circuit“ramctl.” The ram control circuit issues word line select enablesignals, a read command, a write command, and some precharge signals.The ram control circuit generates the signals to control access to andfrom the random access memory “ram.”

FIGS. 7.0302AA-AC provide a circuit drawing of a RAM array “ram 8×4.”Each RAM array is made up of four rows and eight columns of RAM cells.

FIG. 7.030201 provides a circuit drawing of a single RAM cell. In theillustrated embodiment, the RAM cell is a six transistor RAM cell. Fourtransistor RAM cells are employed in alternative embodiments.

FIGS. 7.0303AA-AD provide a circuit drawing of a RAM precharge circuit“rampch.” FIGS. 7.0304AA-AD provide a circuit drawing of a RAM prechargecircuit “ramdch.” In the illustrated embodiment, this circuit has beendisabled as is shown in the figure. The RAM precharge circuits providesprecharge signals to speed up writing to and reading from RAM cells.

FIG. 7.0305 provides a circuit drawing of a RAM address buffer “ramadb.”The RAM address buffer isolates the capacitive load presented by the RAMcircuits from the address bus.

FIGS. 7.0306AA-BA provide a circuit drawing of a RAM word line driver“ramwdr.” The RAM wordline driver is a predecoder. It takes two addressinputs and generates four possible select lines “P0-P3” which are usedin a row decoder (discussed below) for the RAM.

FIGS. 7.0307AA-BB provide a circuit drawing of a RAM word line decoder“ramwdec.” The RAM word line decoder receives the select lines from theRAM wordline driver in conjunction with four other addresses “AD0-AD3”to select a unique word line. A word line is a row of RAM cells withinthe RAM.

FIGS. 7.0308AA-BB provide a circuit drawing of a RAM column selectdecode circuit “ramcdec.” The RAM column select decode circuit usesthree address lines “AD5-AD7” to generate eight select lines“CSEL0-CSEL7.”

FIGS. 7.0309AA-BG provide a circuit drawing of a RAM column selectormultiplexor “ramcsel.” The RAM column selector multiplexor uses theoutput select lines “CSEL0-CSEL7” from the RAM column select decodecircuit “ramcdec” to connect one pair of bit or column lines out ofeight pairs “BIT0N/P”-“BIT7N/P” onto a bus. The bus goes to a sense ampor to a write driver, depending on whether a RAM cell is being read orwritten. There are eight of these RAM column selectors side by side,functioning in the same manner. With any one selection, one of eightpairs are selected on FIGS. 7.0309AA-BG, but there are seven moresimilar selections taking place so an entire byte of RAM is selected atone time.

FIGS. 7.0310AA-BB provide a circuit drawing of a RAM databus interface“ramdb.” The RAM databus interface includes a sense amp and write driverfor the RAM. The RAM databus interface receives the output/input lines“BIT0N/P”-“BIT7N/P” from the RAM column selector “ramcsel.” Selected RAMcells can either be sensed or written.

FIGS. 7.04AA-HJ provide a circuit drawing of a ROM “rom.” The ROM has4096 bytes of contact programmable memory. ROMs of multiple integratedcircuits 16 are simultaneously mass programmed. In the third to the lastmask step, each particular cell of ROM is programmed with a zero or aone. The ROM does not include the information about the lot number,wafer number and die number discussed elsewhere herein. The ROM isprogrammed at the time of manufacture, whereas the information about thelot number and wafer number and die location is stored after themanufacture of the wafer using an electrically programmable or laserfuse programmable, or electrical fuse programmable structure.

FIGS. 7.0401AA-BB provide a circuit drawing of a ROM control logiccircuit “romctl.” The ROM control logic circuit provides signals toallow the contents of eight memory cells of the ROM, one byte to be readout at a time.

FIGS. 7.0402AA-AB provide a circuit drawing of a ROM bit line prechargecircuit “ROMPCH.” The ROM bit line precharge circuit precharges bitlines of the ROM. Bit lines are the vertical lines in the array of ROMcells on which the voltage that is sensed appears after selected ROMcells are accessed.

FIGS. 7.0403AA-BB provide a circuit drawing of a ROM word line driver“romwdr.” The ROM word line driver (or row driver) takes address inputs“A7-A9” and generates enable signals “WDR0-WDR7” to select row lines ofthe ROM.

FIGS. 7.0404AA-DC provide a circuit drawing of a ROM word block decoder“romwdec_rev.” The ROM word block decoder has as inputs the enablesignals “WDR0-WDR7” from the ROM word line driver “romwdr” plus otheraddresses to generate actual word line signals themselves. A word linesignal selects a row of ROM cells.

FIGS. 7.0405AA-BA provide a circuit drawing of a ROM bit line addressdriver “rombldr.” The ROM bit line address driver buffers some of theaddresses so they are capable of driving a large decoder structure“rombldec” (described below).

FIGS. 7.0406AA-CK provide a circuit drawing of a ROM bit line decoder“rombldec.” The ROM bit line decoder provides a decoder structure forselecting a particular ROM bit line out of thirty-two bit lines. Thereare eight such “rombldec” circuits, allowing simultaneous selection ofeight bit lines.

FIGS. 7.0407AA-AB provide a circuit drawing of a ROM sense amplifier“romsns.” The ROM sense amplifier is the sense amp used for determiningthe state of a particular ROM bit being accessed. Eight ROM bit senseamplifiers are used.

FIGS. 7.05AA-CB provide a circuit drawing of an instruction register“insreg.” The code or program that controls the operation of theprocessor is stored in the ROM. The instructions stored in the ROM aretransferred one at a time to this instruction register “insreg” so thatthey can be interpreted and the processor can carry out the operationsrequired by that instruction. After the integrated circuit wakes up, itsoperation is controlled by the wake up and clock recovery circuits.After the integrated circuit locks on to the clock and a valid start(Barker) code is received, the processor turns on and the program storedin the ROM takes over from that point. The program performs functionssuch as determining if the integrated circuit 16 is in a power up cycle.If the device 12 is in a power up cycle, the processor performs varioustasks relevant to power up. If the integrated circuit 16 is receiving acommand from an interrogator, the program will determine which commandand then go through a sequence of required steps in order to respondappropriately to that command. Then the program allows the integratedcircuit 16 to go back to sleep.

FIGS. 7.0501AA-BB provide a circuit drawing of an instruction register“insrcel” included in the instruction register “insreg.”

FIGS. 7.06AA-CN provide a circuit drawing of an instruction decoder PLA“insdec.” The instruction decoder PLA interprets what is in theinstruction register “insreg” and issues all the enable signalsnecessary to effect performance of the functions called for in thatinstruction. Details of the instruction decoder PLA are shown in FIGS.7.0601AA-HI; 7.0602AA-JH; 7.0603AA-JI; and 7.0604AA-JI.

FIGS. 7.0601AA-HI provide a circuit drawing of an instruction decoder(first section) “insdec1.”

FIGS. 7.0602AA-JH provide a circuit drawing of an instruction decoder(second section) “insdec2.”

FIGS. 7.0603AA-JI provide a circuit drawing of an instruction decoder(third section) “insdec3.”

FIGS. 7.0604AA-JI provide a circuit drawing of an instruction decoder(fourth section) “insdec4.” FIG. 7.060401 provides a circuit drawing ofan instruction decoder ROM amp “insramp” included in the circuit ofFIGS. 7.0604AA-JI, 7.0601AA-HI, 7.0602AA-JH, and 7.0603AA-JI. FIG.7.060402 is a circuit drawing of an instruction decoder PLA amp“insramp” included in the circuit of FIGS. 7.0604AA-JI, 7.0601AA-HI,7.0602AA-JH, and 7.0603AA-JI. FIG. 7.060403 is a circuit drawing of aninstruction decoder PLA latch “insplat” included in the circuit of FIGS.7.0604AA-JI, 7.0601AA-HI, 7.0602AA-JH, and 7.0603AA-JI.

FIGS. 7.07AA-BB provide a circuit drawing of a conditional qualifierdecoder “cqualdec.” Certain instructions behave differently depending oncertain conditions (e.g., whether a carry bit is set), and theconditional qualifier decoder looks for these conditions.

FIGS. 7.08AA-CA provide a circuit drawing of a databus latch andprecharge circuit “dblatch.” Data is bused around in eight bit bytes,and the databus latch and precharge circuit drives the databus. The databus is in a precharge high state when the data bus is not being used.Whichever source of data is selected to put its information on the buswill then drive selected bits low if appropriate.

FIGS. 7.09AA-BF provide a circuit drawing of an arithmetic logic unit“alu.” The arithmetic logic unit “alu” is a basic arithmetic logic unitthat provides enough flexibility to perform the functions that areneeded for the RFID task. Details of the arithmetic logic unit areprovided in drawings below.

FIGS. 7.0901AA-CE provide a circuit drawing of an ALU low byte“alubyt1”. There are eight bits within the ALU low byte that are allprocessed simultaneously.

FIGS. 7.090101AA-AD provide a circuit drawing of a ALU bit “alubit1”included in the ALU low byte “alubyt1.” FIGS. 7.090101AA-AD show theregisters contained within each bit of the ALU. The registers include anA cell “aluacell” and a B cell “alubcell” which are the primaryregisters. The data on which arithmetic or logical operations are to beperformed reside typically in the A cell “aluacell” or the B cell“alubcell.” The registers further include a program counter “alupc,” astack pointer “alurcell,” a data pointer “alurcell,” and a memoryaddress register “alumar” that provides for indirect addressing. The ALUbit “alubit1” further includes an adder “aluadd” and a slave register“aluslave” to the adder.

FIG. 7.09010101 is a circuit drawing showing details of construction ofan ALU bit decoder cell “alubdec” included in the ALU bit.

FIG. 7.09010102 is a circuit drawing showing details of construction ofthe ALU B register cell “alubcell” included in the ALU bit.

FIG. 7.09010103 is a circuit drawing showing details of construction ofthe ALU A register cell “alubacell” included in the ALU bit.

FIG. 7.09010104 is a circuit drawing showing details of construction ofthe ALU program counter “alupc” included in the ALU bit.

FIG. 7.09010105 is a circuit drawing showing details of construction ofthe ALU register cell “alurcell.” Such cells are used for a stackpointer, data pointer, etc.

FIG. 7.09010106 is a circuit drawing showing details of construction ofthe ALU memory address register “alumar” included in the ALU bit.

FIG. 7.09010107 is a circuit drawing showing details of construction ofthe ALU slave cell “aluslave” for the ALU adder “aluadd.”

FIG. 7.09010108 is a circuit drawing showing details of construction ofthe ALU adder “aluadd” included in the ALU bit.

FIGS. 7.0902AA-BD provide a circuit drawing for an ALU high byte“alubyth” which functions similarly to the ALU low byte “alubyt1.” TwoALU bytes are provided so that sixteen bit commands can be processed.

FIGS. 7.090201AA-AC provide a circuit drawing of a bit “alubith”included in the ALU high byte “alubyth.”

Details of Low Power Dormant Mode

It is sometimes desirable to prevent the integrated circuit 16 fromresponding to commands from an interrogator. For example, aftercommunication with a particular device 12, it is sometimes desirable toprevent that particular device 12 from responding to a subsequentinterrogation that is intended for a different device 12. If, forexample, the device 12 is used in connection with an access gate, afteran interrogator has read a badge containing the device 12 as acontrolled access point is passed, the interrogator no longer has a needto communicate with that badge. The interrogator instead would want topick up subsequent badges passing through the access gate. In addition,when the interrogator no longer has a need to communicate with aparticular device 12, it is desirable that the device 12 stay in thesleep mode to conserve battery power.

In one embodiment, the device 12 is put into an unresponsive state byusing a counter which is set to a desired time via a radio frequencycommand. The device will then not respond to Identify commands(described below in greater detail) used by an interrogator to requestinformation from a device 12. In this embodiment, the unresponsive statecan be cancelled by a radio frequency command. However, this embodimentis disadvantageous in that the device must wake up to process incomingcommands and abort if the command is an Identify command. This consumesbattery capacity.

In a preferred embodiment, the device 12 can be placed in a dormant modevia a radio frequency command. The dormant mode cannot be cancelled.When in the dormant mode, the device 12 does not wake up to look forincoming commands.

FIGS. 7.10AA-CC provide a circuit drawing of a timed lock out divider“tld.” The timed lock out divider takes as an input the low power clockwhich is the same clock that sets the wake up interval for theintegrated circuit 16. The timed lock out divider provides twofunctions. The timed lockout divider provides an alarm timer function,and provides a timed lockout function which is used for the dormant modefunction and for the timed lockout of Identify commands.

The alarm timer is set to go off in intervals, such as about every oneminute. As an alarm timer, the timed lock out divider causes theintegrated circuit 16 to wake up and check for threshold violations inalarm mode. Such threshold violations would be triggered by analogsensors such as temperature sensors, magnetic sensors, etc.

The timed lock out divider also allows, by RF command from aninterrogator, a user to disable a device 12 to make it not respond for aprescribed period of time (i.e., allows the user to place the device 12in the dormant mode). The prescribed period of time can be set invarious increments. For example, in the illustrated embodiment, theincrements are one second increments from one up to 255 seconds.

When in the dormant mode, the device 12 does not periodically switch tothe receiver on mode to check for the presence of radio frequencycommands. Therefore, power is conserved.

This dormant mode function is useful for the same reasons that thecancellable timed disabling is useful. If, for example, the device 12 isused in connection with an access gate, after an interrogator has read abadge containing the device 12 as a controlled access point is passed,the interrogator no longer has a need to communicate with that badge.The interrogator instead would want to pick up subsequent badges.Therefore, the interrogator can instruct the device 12 to not respondfor a certain time, so as to prevent an unwanted response of a device12, after having communicated with that device 12, but with increasedpower savings over the cancellable timed disabling. Because wake ups aredisabled, current consumed by the device 12 is very low; e.g., 1 μA.

FIG. 7.1001 provides a circuit drawing showing details of constructionof a timed lock out divider cell “tldcel” included in the timed lockoutdivider “tld.”

FIGS. 7.11AA-AB provide a circuit drawing of a timed lock out register“tloreg.” This register acts as a down counter and is selectively setwith the desired lockout time, from 1 to 255 seconds.

FIGS. 7.1101AA-AC provide a circuit drawing of a timed lock out registercell “tlorcel” included in the timed lockout register.

FIGS. 7.12AA-AC provide a circuit drawing of an read/write controlregister or output register “oreg.” There are a number of these outputregisters. The output registers allow the processor to send controlsignals out to various peripheral circuits to cause them to functionwhen required.

FIG. 7.1201 provides construction details of a control register cell“regcell” included in the output register “oreg.”

FIGS. 7.13AA-BA provide a circuit drawing of a status register “sreg.”The processor uses the status register to monitor the status of linessupplied from various blocks of circuitry.

FIGS. 7.1301AA-AB provide a circuit drawing of a status register cell“sregcel” included in the status register.

FIGS. 7.14AA-AB provide a circuit drawing of a serial input output block“sio.” The serial input output circuitry is the data path for datareceived and for the data to be transmitted. This circuit controls thetransfer of the serial stream of data received from the receiver intothe processor. The circuit also controls the transfer of the transmitserial data stream from the processor out to the transmitter. The serialinput output circuitry comprises two blocks: a block “siodata” thatprocesses data, and a controller “sioct1” that runs the block thatprocesses data.

FIGS. 7.1401AA-AB provide a circuit drawing of a serial input outputdata path “siodata.” When the integrated circuit 16 is in a transmitmode, data enters the bit registers “sioreg” from the top of the figure,and the data is transferred down to the registers “siobdlat” and“siobdlat_inv” which are the row of blocks second up from the bottom ofthe figure. The intermediate stages “sioxor” are all exclusive or gatesthat are used to generate check bits according to the previouslymentioned modified Hamming code. The extra five bits “P0-P4” appended tothe eight data bits “D0-D7” are generated by the exclusive-or gates, andthen all thirteen bits are transferred to the registers “sioshr” whichare the row of blocks at the bottom of the figure. The thirteen bits areserially shifted out to the right of the figure.

When the integrated circuit 16 is in a receive mode, a reverse sequencetakes place. Data is shifted into the thirteen bit registers “sioshr”shown on the bottom of the figure, then transferred up to the registers“siodblat” immediately above the shift registers “sioshr” in the figure.Then the exclusive or circuitry “sioxor” uses the data and the checkbits to determine whether there are any errors. If there are anycorrectable errors, they are corrected at that point. The serial inputoutput data path “siodata” can also detect double bit errors which arenot correctable. If a double bit error is detected, a signal is providedat the upper left of the figure to the processor that an uncorrectableerror has occurred. Assuming that there is no uncorrectable error, theeight corrected bits are now present as inputs to the top row ofregisters “sioreg.” The eight corrected bits are then transferred in tothe top row of registers. From the top row of registers “sioreg,” thecorrected bits are transferred in parallel to the processor.

FIGS. 7.140101AA-AB provide construction details of the serial inputoutput register cell “sioreg” included in the serial input output datapath “siodata.”

FIGS. 7.140102AA-GF provide construction details of the serial inputoutput exclusive or circuit “sioxor” included in the serial input outputdata path “siodata.”

FIGS. 7.140103AA-AB provide construction details of the bidirectionallatch “siobdlat_inv” included in the serial input output data path“siodata.”

FIGS. 7.140104AA-BB provide construction details of the shift register“sioshr” included in the serial input output data path “siodata.”

FIGS. 7.140105AA-AB provide construction details of the bidirectionallatch “siobdlat” included in the serial input output data path“siodata.”

FIGS. 7.1402BA-EI provide a circuit drawing of the previously mentionedcontrol logic “sioct1.” The control logic “sioct1” generates all theclocking and the signals that control when data is transferred fromregister to register.

FIGS. 7.140201AA-BB provide a circuit drawing showing constructiondetails of the counter bit “siocbit” included in the control logic“sioct1.”

FIGS. 7.15AA-EC provide a circuit drawing of a data interleaver “dil.”In a number of modulation schemes used or selectively used by theintegrated circuit 16, differential encoding is employed. Use ofdifferential encoding in the integrated circuit 16 makes possible asimpler receiver in the interrogator. However, if an error occurs in theprocess of differential encoding, it necessarily corrupts two adjacentbits. The modified Hamming code cannot correct errors where two adjacentbits are in error. This problem is solved by interleaving two bytes. Bitby bit, the first bit of one byte is shuffled next to the first bit ofanother byte and so on through all thirteen bits. This way, whendifferential encoding is performed, which may possibly create twoadjacent errors, the two bytes are deinterleaved and separated at thereceiver so that the bytes are in separate error corrective words. Theerrors can then be fixed.

The data interleaver works by shifting data in from a data input“SIOTXD” (on the upper left of FIGS. 7.15AA-EC). Twenty-six bits areshifted into the registers “dil_sreg” shown along the top of FIGS.7.15AA-EC, then all twenty-six bits are simultaneously shifted to thelower registers “dil_plsreg” and scrambled in order simultaneously bywiring interconnections between the registers “dil_sreg” and theregisters “dil_plsreg” shown in FIGS. 7.15AA-CC. Thus, a new interleaveorder is generated on transfer from the registers “dil sreg” to theregisters “dil_plsreg.” Then, the contents of the registers “dil_plsreg”are shifted out (to the right in the view of FIGS. 7.15AA-EC) in aserial, bit by bit fashion, through line “DILTXD.”

FIGS. 7.1501AA-CA provide a circuit drawing showing construction detailsof the shift register “dil_sreg” included in the data interleaver “dil.”

FIGS. 7.1502AA-CA provide a circuit drawing showing construction detailsof the parallel load shift register “dil_plsreg” included in the datainterleaver “dil.”

FIG. 7.150201 provides a circuit drawing showing construction details ofa shift register bit “dil_sregbit” included in the parallel load shiftregister “dil_sregbit” and in the shift register “dil_sreg.”

FIGS. 7.16AA-CD provide a circuit drawing of a convolutional encoder“conv.” In the illustrated embodiment, convolutional encoding isdisabled. However, in one embodiment, convolution encoding is provided.The circuitry of FIGS. 7.16AA-CD performs more functions than justconvolutional encoding. The circuitry of FIGS. 7.16AA-CD also includes apreamble generator. In one embodiment, a series of zeros are generatedas a preamble. However, in the illustrated embodiment, a pattern ofalternating zeros and ones (0101) is generated for DPSK backscatter. Thecircuitry of FIGS. 7.16AA-CD also includes a clock for the SIO “sio.”

FIG. 7.1601 provides a circuit drawing showing construction details of ashift register cell “convshr” included in the convolutional encoder“conv.”

FIG. 7.1602 provides a circuit drawing showing construction details of asummer “convsum” included in the convolutional encoder “conv.”

FIGS. 7.17AA-BB provide a circuit drawing of a shift register datamultiplexor “shdcel.” The shift register data multiplexor provides aport into the processor. It does a selection among eight sources on theintegrated circuit 16, and connects only one of them for shifting ofdata for transfer into the A register.

FIGS. 7.18AA-CC provide a circuit drawing of a digital port outputcontroller “doutport.” The device selectively reads data via a digitalport in response to a radio frequency command, instead of by radiofrequency reception, and the device selectively writes data via adigital port in response to a radio frequency command, instead of byradio frequency. The digital port output controller circuit controlsthese functions. The digital port output controller circuit alsoincludes a clock in order to synchronize the transfer of the data ineither direction (input or output).

The RF processor “rfproc” shown in FIGS. 6AA-EK contains the receiver30, the transmitter 32, the clock recovery and data recovery circuit 38,and the wake up timer and logic circuit 36. The RF processor “rfproc” isshown in greater detail in FIGS. 8AA-CB.

FIGS. 8AA-CB provide a circuit drawing of a RF processor “rfproc.” TheRF processor “rfproc” includes a receiver “rx” (which is the receiver 30of FIGS. 6AA-EK), a transmitter “tx” (which is the transmitter 32 ofFIGS. 6AA-EK), a low power frequency locked loop “lpfll,” a counter bit“lpfll_cbit,” a receiver wake up controller “rxwu” (which is the wake uptimer and logic circuit 36 of FIGS. 6AA-EK), and a digital clock anddata recovery circuit “dcr” (which is the clock and data recoverycircuit 38 of FIGS. 6AA-EK). Thus, RF processor “rfproc” includes theclock that sets the wake up interval, as well as logic that performstests on the incoming signal to see whether the incoming signal is avalid signal such that the integrated circuit 16 should stay awake.

FIGS. 8.01AA-DE provide a circuit drawing of the receiver “rx” includedin the RF processor. In the illustrated embodiment, the receiver “rx”includes a Schottky diode detector “diodedet.” In the illustratedembodiment, the Schottky diode detector “diodet” is an inductorlessSchottky diode detector. Instead of employing inductors in the diodedetector to supply bias current to the diode, the diode detectorincludes a current source which drives current through both an antennaand a Schottky diode included in the detector. The inductorless Schottkydiode detector is described in more detail below. FIGS. 8.01AA-DE alsoillustrate a CMOS detector “cmosdet” that is used in accordance with analternative embodiment, but which is not used in the illustratedembodiment. The output of the Schottky diode detector is applied througha series of AC coupled amplifiers. More particularly, in the illustratedembodiment, the Schottky diode detector is applied through amplifiers“videoamp1,” “videoamp2” replicated four times, and then into acomparator. The function of the comparator is to put out a full digitalsignal. The output of the comparator is a base band digitalrepresentation of the command that was sent by the interrogator.

A base band signal is a signal without a carrier frequency present. Theoutput of the comparator is a signal that is the equivalent of thesignal that was used to modulate the carrier back at the interrogator.

The receiver “rx” includes a RF detect circuit “rxdetect.” The RF detectcircuit determines when a modulated radio frequency signal is present atthe receiver and the output of the receiver is switching between highand low states. The receiver “rx” includes a bias block “rxbias” thatprovides currents to the various amplifiers “videoamp1,” “videoamp2,”etc. The receiver “rx” further includes logic that bypasses the receiverwhen a user selects not to use RF for an input, but rather to provide abase band input signal directly in digital form. The user may make theselection to bypass the receiver, for example, for testing or exercisingthe integrated circuit 16. The user may also make the selection inapplications where the receiver portion of the chip is not required, butthe integrated circuit 16 is used to transmit information (e.g., forperiodic transmissions).

Schottky Diode RFID Detector

Overview

For purposes of realizing a cost effective and low power radio frequencyreceiver on an RFID tag, a simple Schottky diode receiver is utilized.The receiver is formed from a Schottky diode detector, an amplifier, andthe receiving antenna “rxantenna”. With the implementation of a singleintegrated circuit 16 RFID tag, an easy and low cost technique forconfiguring the frequency of operation on a tag is needed. Receiverfrequency characteristics can be tailored by selecting an appropriatelysized antenna to be coupled to the integrated circuit 16 that supportsthe Schottky diode detector. Furthermore, adjustment of bias currentacross the Schottky diode can be used to realize a desired resistancethere across, enabling tuning or detuning of the receiver.

For purposes of enabling simplified representation, FIG. 29 illustratesa simplified circuit schematic for one embodiment of a receiver 80having a Schottky diode detector 84 and antenna 44.

The detector 84 includes a Schottky diode 86 having an anode connectedto the antenna 44 and having a cathode.

The exemplary antenna 44 is formed from a loop or folded dipoleconstruction. The antenna 44 performs band pass filtering.

The detector 84 further includes an ideal current source 88 connected tothe cathode of the Schottky diode 86 and driving current through theantenna and Schottky diode 86 in the direction from the anode to thecathode. The current source 88 is an ideal current source, and isconfigured to forward bias the Schottky diode 86, realizing a desiredresistance (or impedance) in the process.

The detector 84 further includes a capacitor 90 connected between thecathode of the Schottky diode 86 and ground. The capacitor 90 provides aradio frequency short to ground so that all radio frequency voltageappears across the Schottky diode 86. This maximizes a base band signalproduced by the Schottky diode 86.

The detector 84 further includes a capacitor 92 having a first terminalconnected to the cathode and having a second terminal defining an outputof the detector 84. The capacitor 92 provides an AC short to videofrequency, and defines the output of the detector 84. The capacitor 92allows different bias levels in the detector and at the input of a videoamplifier connected to the output of the detector 84. Details of theactual circuit implementation on integrated circuit 16 (of FIG. 5) willbe discussed below with reference to FIGS. 5, 8AA-CB, 8.01AA-DE,8.0101AA-CB, 28, 29, and 30.

Antenna Implementation

Preferably, the antenna “rxantenna” is constructed and arranged to forma folded dipole antenna, consisting of a continuous conductive path, orloop of microstrip. The terminal ends of the antenna 44 loop each form aconductive lead that electrically interconnects with the integratedcircuit 16 of FIG. 5. According to the actual circuit layout of FIG. 6,antenna “rxantenna” is connected to the integrated circuit 16 via theexposed conductive bonding pad labeled “rxantenna—Pad D”. Alternatively,the antenna can be constructed as a continuous loop antenna. In thiscase, the antenna is constructed from a continuous piece of conductivemicrostrip configured in the shape of a square or circle to form a loopantenna.

In assembly, antenna 44 (as well as antenna 46) is depicted inelectrically conductive and bonded relationship with “rxantenna—Pad D,”shown on the integrated circuit 16 of FIG. 6. Similarly, antenna 46 isbonded to “txantenna—Pad AA”. The preferred assembly technique,discussed below, involves a flip-chip epoxy bonding technique whereinthe antennas 44 and 46 of FIG. 5 are actually printed onto the back faceof the plastic card or carrier (e.g. card 11 of FIG. 2 and stamp 20 ofFIG. 3), after which the integrated circuit 16 is bonded to the antenna,as well as to the battery, using a conductive epoxy.

Preferably, the antennas 44 and 46 are printed onto the back side of thecard or stamp, forming each microstrip loop antenna thereon. Forexample, the antenna can be silk screened onto the card with aconductive polymer thick film. Alternatively, a conductive silver filledepoxy can be used. Alternatively, the antenna can be formed from aseparate piece of conductive material, for example, from a piece of wireor conductive ribbon that is glued to the back of the card.

One exemplary technique for assembling the postage stamp 20 of FIG. 4 isprovided here below. The same technique can be used to assemble thebadge 10 of FIG. 2, or any other similarly constructed tag having arigid support or substrate similar to plastic cards 11 and 21. First,antennas 44 and 46 (of FIG. 5) are mounted to a back face of the card.Preferably, the above elements are simultaneously printed onto the backof a large sheet of plastic with a conductive silver printed thick film,after which the cards are individually separated, or cut from the sheet.Pads on the integrated circuit 16 form enlarged connection points forelectrically bonding each antenna 44 and 46 to “rxantenna—Pad D” and“txantenna—Pad AA” of FIG. 6 and for connections to a power supply.Next, the card is positioned front face down onto a rigid support plate.Then integrated circuit 16 (of FIG. 4) is mounted to the pads withconductive beads of epoxy. Finally, the battery 18 is bonded along itsbottom face with a bead of conductive epoxy, after which conductiveepoxy is used to electrically connect the opposite terminal or top ofthe battery with a corresponding conductive die pad.

Subsequently, a metal dam sized to conform generally to the outerperipheral shape of the card 20 is placed over the back of the card. Thedam functions as an outer template while a thin layer of non, conductiveepoxy (not shown) is applied to the back of the card 20, preferablyhermetically sealing in the integrated circuit 16, antenna and battery.Preferably, the thin coat of epoxy consists of a coating, barely sothickenough to cover over the components forming the device. One benefitprovided by this construction technique is the elimination of anyvisible bumps in the tag which can result when constructing the tag byheat sealing two or more pieces of plastic card together to trap thedevice 12 therein. However, a lesser preferred construction of thisinvention envisions forming the tag, e.g. badge 10, stamp 20, or someother tag, with such a heat sealed sandwich of plastic cards.

Preferably, the above technique for mounting integrated circuit 16 tocard 20 (of FIG. 4) consists of a flip-chip mounting technique. Oneexample of a flip-chip mounting technique is disclosed in pending U.S.patent application Ser. No. 08/166,747, “Process of Manufacturing anElectrical Bonding Interconnect Having a Metal Bond Pad Portion andHaving a Conductive Epoxy Portion Comprising an Oxide Reducing Agent,”listing Rick Lake and Mark E. Tuttle as inventors, now U.S. Pat. No.5,480,836 and incorporated herein by reference.

Integrated Circuit Implementation

According to FIG. 8, the Schottky diode detector “diodet” is configuredwithin receiver “rx” to receive radio frequency signals via receivingantenna “rxantenna”. One exemplary receiving antenna configuration isdepicted in FIG. 5, denoted generally by reference numeral 44. Inoperation, the Schottky diode detector and the receiving antennacooperate to form a tunable receiving circuit. Signals detected by theSchottky diode detector are input to a five stage amplifier, then acomparator, for further signal conditioning. The output of thecomparator is a digital representation of the received baseband signal.

As shown in FIG. 8, receiver “rx” is an Amplitude Shift Keying (ASK)receiver. This is also known as an AM receiver. The illustratedembodiment employs On Off keying (OOK) wherein a digital one (“1”) isrepresented by the presence of the RF carrier, and wherein a digitalzero (“0”) is represented by the absence of the carrier.

FIGS. 8.01AA-DE illustrate in greater detail the circuit implementationof receiver “rx”. According to this embodiment, Schottky diode detector“diodedet” receives input signals via an input “rxantenna,” and biasvoltages for the current source via a pair of inputs “bias1” and“bias2”. A pair of output signals “OUTN” and “OUTP” leave “diodedet” forinput to a serially connected chain of amplifiers, and a comparator. Thearray of amplifiers comprise five video amplifiers, labeled “videoamp1”and “videoamp2”. Bias voltages are applied to “bias1” and “bias2” via“rxbias,” a bias circuit which generates all bias voltages required bythe receiver.

A circuit “rxdet” shown in FIGS. 8.01AA-DE receives the output signalfrom the comparator, via combinational logic, with “digrxdata” and“digrx”. The output signal “RFDET” is driven high if there is a signalat the output of the comparator. The resulting signal input into “rxdet”is received via “dataIn.” Additional inputs to “rxdet” include“lowrate,” “Vref,” “Vbias1,” “Vbias2,” “Vreg,” and “enable.” Furtherdetails of “rxdet” are disclosed below with reference to FIGS.8.0106AA-CD, entitled “RF Detect”.

FIGS. 8.0101AA-CB illustrate one embodiment for realizing the Schottkydiode detector “diodedet” of FIGS. 8.01AA-DE. Namely, a Schottky diodeis forward biased from receiving antenna, coupled at “ANT,” to adetector output “OUTP.” A second Schottky diode is forward biased fromVdd to a detector output “OUTN.” Two current sources are formed by fourtransistors, and are driven by bias voltages at “bias1” and “bias2.” Acapacitor is coupled to Vss, between each Schottky diode and associatedoutput, “OUTP” and “OUTN,” respectively. Furthermore, an array ofparallel capacitors are provided in series between each Schottky diodeand associated output, “OUTP” and “OUTN,” respectively. The array ofparallel capacitors acts as a single capacitor. Each Schottky diode isformed from an array of Schottky diodes. In order to use standardcontact hole sizes, each Schottky diode is formed from an array ofSchottky diodes connected together in parallel to act as a singleSchottky diode.

According to FIGS. 8.01AA-DE, “OUTP” and “OUTN” are input into amultiple (e.g., five) stage amplifier and into a comparator. In order toavoid amplification of substrate noise, a differential amplifier isemployed for each stage of the multiple stage amplifier. Noise appearsequally on both inputs of each differential amplifier, and, the commonmode rejection of the differential amplifiers impedes transmission ofsubstrate noise. The differential amplifiers amplify a received basebandsignal up to a digital level. A dummy Schottky diode (the lower Schottkydiode in the figures) is connected to the second input of the firstdifferential amplifier.

Details of Realization of Wide Carrier Frequency Bandwidth

In order to meet the wide range of intended applications, it isdesirable to construct the integrated circuit for an RFID tag to realizeoperation of a wide range of carrier frequencies. For example, severaldesirable carrier frequencies for the device disclosed in FIGS. 5 and 6are 915, 2450, and 5800 MegaHertz bands. Frequency selectivity isrealized in the device of FIGS. 5 and 6 by appropriately configuringexternal antennas and internal circuit components of the integratedcircuit. For the case of a single integrated circuit with an activeonboard transmitter, it is necessary to design circuit components intothe integrated circuitry prior to mounting and encapsulation of theintegrated circuit with an antenna inside of a package. Hence, thecircuit components needed to facilitate tailoring of the carrierfrequency must be “designed in” the integrated circuit. In the case of abackscatter transmitter, components included in the integrated circuitcan be selected so as to allow operation over a wide range of carrierfrequencies, the selection being made by choice of antenna.

According to FIGS. 8.01AA-DE, the number of amplifiers that need to beimplemented via “videoamp1” and “videoamp2” is determined based upon themagnitude of the minimum detected signal and the required osignal tonoise (S/N) ratio. Amplification is sufficient to produce full digitallevels at the output of the comparator.

The capacitor configured to ground in the video receiver circuit of FIG.29 (and FIGS. 8.0101AA-CB) is used to separate the radio frequency (RF)from the “VIDEO AMP” side of the video receiver circuit. The capacitoris sized to impart an effective short circuit to ground at radiofrequency, thereby ensuring that all of the radio frequency (RF) voltageappears across the Schottky diode terminals. Additionally, the capacitorshould be sized small enough at video frequencies, so that the capacitordoes not load down the video amp circuit.

The capacitor configured in series in the video receiver circuit of FIG.29 (and FIGS. 8.0101AA-CB) is used to block out the DC component of avoltage to “VIDEO AMP” while retaining the AC component. In this manner,the series capacitor forms a “blocking” capacitor or “coupling”capacitor.

Details of Inductorless RF Detector

A second desirable feature for the integrated circuit of an RFID tag isto eliminate the need to use inductors when constructing the Schottkydiode detector. One technique for providing a bias current to a Schottkydiode is disclosed in FIG. 28. FIG. 28 illustrates a receiver 60including an antenna 62 and a Schottky diode detector 64. The receiver60 includes inductors 68 and 70 used to provide the bias current viavoltage source 74 with this implementation. A capacitor 76 is shunted toground, and a second capacitor 78 is placed in series, providing ACcoupling to the video amplifier. Several variations of suchinductor-based bias current implementations are described in a paperentitled “Designing Detectors for RF/ID Tags,” by Raymond W. Waugh ofHewlett-Packard Company, submitted for presentation at the RF Expo, SanDiego, Feb. 1, 1995, and which is already incorporated by reference.Inductors are required in all of these constructions, but theirimplementation on an integrated circuit proves difficult because ofproblems inherent in forming inductors in an integrated circuit. Thecircuit in FIG. 29 eliminates the inductors by biasing the Schottkydiode with a high impedance current source. A current sink is providedby connecting the far end of the antenna to Vdd.

Details of Elimination of Overdrive Problem

The Schottky diode detector circuit implementation of FIGS. 8.0101AA-CBrealizes a technique for negating the effect of high power radiofrequency (RF) input levels on the Schottky diode detector. Moreparticularly, when high level radio frequency (RF) power is present atthe antenna “rxantenna,” e.g. when the RFID tag antenna is close to thetransmitting antenna of an interrogator, the signal present on node “A”of FIG. 29 becomes large. For example, the signal on node “A” could beseveral hundreds of millivolts. The rising and falling edges of thedetected signal are controlled by two separate time constants which arevery different, as shown in FIG. 31. As shown by the high power signalof FIG. 31, the high power signal has a rising edge which is fast, orhas a very steep, nearly vertical slope. The nearly vertical slope ofthe rising edge results because the rising edge is controlled by theeffective resistance of the Schottky diode (about 1 kOhm) multiplied bythe capacitance of capacitor 90 (Crf) (about 1-10 pF). Hence, theresulting time constant is about 1 to 10 nanoseconds.

In contrast, the falling edge of the detected signal in FIG. 31 iscontrolled by the c urrent source 88 as it discharges capacitor 90(Crf), which takes approximately 100 nanoseconds. As a result, thevoltage waveform at node “A” is distorted. According to the amplifieddigital version of the signal, shown in FIG. 31, the signal in the highpower case is distorted by the unequal rise and fall times. The onset ofeach fall for the digital version is triggered at the cross-over point,which deviates substantially from that of the low power signal. Such adistortion poses a serious problem for implementing clock recoveryschemes, which rely on accurate edge-to-edge timing.

To overcome the above-mentioned problem, the integrated circuit 16 ofFIG. 6 uses only rising edges for clock recovery. Hence, the distortedfalling edges are avoided altogether. As becomes apparent from viewingthe amplified digital signal of FIG. 31, rising edge to rising edgetiming is not affected by the slow falling edges. Therefore, the clockcan be accurately recovered.

FIG. 30 illustrates a circuit 93 including a Schottky diode detector 94,and an antenna 44 connected to the Schottky diode detector 94. Moreparticularly, in the illustrated embodiment, the Schottky diode detector94 includes a Schottky diode 96 having a cathode connected to theantenna 44 and an anode. The Schottky diode detector 94 further includesa current source 98 driving current in the direction from the anode tothe cathode of the Schottky diode 96 and through the antenna 44. TheSchottky diode detector 94 further includes a capacitor 100 connectedbetween the anode of the Schottky diode 96 and ground; and a capacitor102 connected between the anode of the Schottky diode 96 and an outputof the diode detector 94 which is connected to an amplification circuit.The same technique used with respect to FIG. 29 can also be implementedfor the Schottky diode detector circuit of FIG. 30. However, for thiscase, only the rising edges are significantly distorted, since theSchottky diode is reversed in direction. Therefore, only the fallingedges are used in clock recovery.

Details of Method of Forming an IC Schottky Structure

A method of forming a Schottky structure that can be employed tomanufacture the Schottky diode detector will now be described. ASchottky diode is a diode in which a metal and a semiconductor form a pnjunction. Electrons injected into the metal have a higher energy levelthan the charge carriers in a semiconductor, and energy storage at thejunction is low because current flow is not accompanied by holemovement.

One embodiment of the invention comprises a Schottky diode 220 includingan n+ region 222 generally encircling or surrounding an n-well region224 (FIG. 41). In the illustrated embodiment, the n+ region 222 isheavily doped; e.g., 1×10¹⁸ atoms/cm³ or greater of n-type material, andthe n-well region 224 is lightly doped; e.g., 1×10¹⁷ atoms/cm³ or lowerof n-type material. The n-well region 224 defines a contact area 226,and the n+ region 222 provides a low resistance interconnect to theSchottky diode 220. The n+ region 222 has a diffused edge 228, and then-well region has a contact edge 230. The distance from the n+ regiondiffused edge 228 to the n-well region contact edge 230 is minimized. Inone embodiment, the distance from the n+ region diffused edge 228 to then-well region contact edge 230 is less than twenty micrometers. In amore preferred embodiment, the distance from the n+ region diffused edge228 to the n-well region contact edge 230 is about two micrometers.

More particularly, the integrated circuit 16 includes a grid pattern ofn+ regions 222. Each region 222 generally encircles or surroundsisolated n-well regions 224 of a large common n-well region 232 underthe n+ regions 222 (FIGS. 41 and 42). This provides for parallelconnection of a selectable number of Schottky diodes 220. As describedelsewhere, the parallel connection of Schottky diodes 220 acts a singleSchottky diode, and allows use of standard sized contact holes. Thenumber of Schottky diodes 220 connected together is selectable to tailorresistance, parasitic capacitance, and electrostatic dischargesensitivity for a specific application.

To form the grid of Schottky diodes 220, the following process steps areperformed.

First, a p-substrate 234 is provided (FIG. 38). Next, n-well region 232is defined relative the substrate 234. Next, an insulator 236 is formedover the n-well region. In one embodiment, the insulator 236 isborophosphosilicate glass (BPSG).

Next, a removal or etching step is performed to remove areas of theinsulator 236 for definition of contact holes 238, and areas 240generally encircling or surrounding the contact holes 238 (FIG. 39). Thecontact holes 238 are not necessarily circular in cross-section; anycross-sectional shape is possible. Similarly, any cross-sectional shapeis possible for the areas 240 surrounding the contact holes 238. In apreferred embodiment, the contact holes 238 all have the same diameter(or peripheral extent) to facilitate subsequent filling of the contactholes 238 (described below in greater detail). In an alternativeembodiment, different contact holes 238 have different sizes. In theprocess of the illustrated embodiment, the contact holes 238 do not needto be completely filled with a conductor, and all contact holestherefore do not need to be the same size.

In the illustrated embodiment, the n+ regions 222 are formed in then-well region 232 by diffusion after the etching has been performed, viathe openings 240 surrounding the contact holes 238. The n+ regions 222can be formed by other processes or in other sequences. For example, then+ regions 222 can be formed before the insulator 236 is formed over then-well region 232.

Next, a Schottky forming metal 242 such as titanium is formed in thecontact hole openings. In the illustrated embodiment, the Schottky|forming metal is deposited on the surface of n-well regions 224 via thecontact hole openings 238. In one embodiment, the thickness of thedeposited metal is about 200 Å. The metal is annealed to form a stablesilicide interface to the n-well silicon.

If it is desired to fill the contact holes, a material such as tungsten246 may be deposited into the contact holes (FIG. 40). The tungsten isthen planarized to form final contact structures (FIG. 41).

Then, an interconnect metallization step is performed (FIG. 42). Forexample, copper doped aluminum 248 is deposited (e.g. sputtered) overthe wafer, then the wafer is masked and etched to remove unwanted areas.The mask defines a pattern to interconnect the contacts as desired.

A variable number of Schottky diodes may be connected in parallel bysimply changing the metal masks and interconnecting only the number ofSchottky diodes required by a particular circuit application. In oneillustrated embodiment, an array of twelve by twelve Schottky diodes(144 total Schottky diodes) is provided (FIG. 43). In another embodiment(FIG. 42), less than all available Schottky diodes are connectedtogether. In one embodiment, only a six by six array (36 Schottkydiodes) is connected together in parallel.

In one alternative embodiment, aluminum is employed instead of tungstenand silicide. In another alternative embodiment, tungsten is employedinstead of aluminum to interconnect contacts, and the step of formingtungsten plugs is omitted.

In an alternative embodiment (FIG. 47), each Schottky diode includes ap+ region 252 encircling a “p-” p-well region 254 and is formed by amethod substantially identical to the method described above except withp-type material substituted for n-type material and vice versa. Moreparticularly, in this alternative embodiment, the following steps areperformed.

First, an n-type substrate 256 is provided (FIG. 44). Next, a commonp-well region 258 is defined relative the substrate 256. The commonp-well region 258 defines the p-well regions 254 for each of theSchottky diodes. Next, an insulator 260 such as borophosphosilicateglass (BPSG) is formed over the p-well region 258. Next, an etching stepis performed to etch away regions of the insulator for definition ofcontact holes, and areas 264 generally encircling or surrounding thecontact holes (FIG. 45). In a preferred embodiment, the contact holes262 all have the same diameter (or peripheral extent) to facilitatesubsequent filling of the contact holes 262 with Tungsten or anotherconductor. In an alternative embodiment, different contact holes 262have different diameters. In the process of the illustrated embodiment,the contact holes do not need to be completely filled, and all contactholes therefore do not need to be the same size.

In the illustrated embodiment, the p+ regions 252 are formed in thep-well regions by diffusion after the etching has been performed, viathe openings 264 encircling the contact holes. The p+ regions 252 can beformed by other processes or at other times. For example, the p+ regions252 can be formed before the insulator is formed over the p-well region258.

Next, a Schottky forming metal 266 such as Titanium is formed in thecontact hole openings 262. In the illustrated embodiment, the Schottkyforming metal 266 is deposited on the surface of the p-well region 258via the contact hole openings 262. In one embodiment, the thickness ofthe deposited metal is about 200 Å. The metal 266 is annealed to form astable silicide interface 268 to the p-well region 258.

If it is desired to fill the contact holes 262, a metal such as tungsten270 is deposited into the contact holes (FIG. 46). The tungsten 270 isthen planarized to form final contact structures.

Then, an interconnect metallization step is performed (FIG. 47). Forexample, copper doped aluminum 272 is deposited (e.g. sputtered) overthe wafer, then the wafer is masked and etched to remove unwanted areas.The mask defines a pattern to interconnect the contacts as desired.

The above described processes for forming a Schottky diode are preferredover a process wherein, after the contact holes a etched, ionimplantation of phosphorus into the holes is performed. In such aprocess, the implant would be a two step process, with a low energyimplant (e.g., 35 keV of 4×10¹² ions/cm²) followed by a high energyimplant (e.g., 120 keV of 4×10¹² ions/cm². Such implants cause a highdoping level at the bottom of the contact hole, which prevents formationof a low leakage Schottky diode. The preferred processes described aboveeliminates these two contact implants, and allows for formation of agood quality Schottky diode.

FIGS. 8.0101AA-CB provide a circuit drawing of the Schottky diodedetector “diodedet.” FIGS. 8.0101AA-CB actually show two Schottky diodedetectors. The lower Schottky diode detector shown in FIGS. 8.0101AA-CBis a replicated or dummy detector which generates a signal for thecompliment side of the differential amplifier “videoamp1.” The structureof the dummy Schottky diode detector is similar to the real Schottkydiode detector so that any noise coupled through ground or possiblythrough Vdd is replicated on both sides of the differential amplifier“videoamp1” and so that the common mode rejection of the amplifier willresult in little noise making it through the amplifier chain. Biascurrent to the Schottky diode detector is provided by the current sourcetransistors having gates connected to “BIAS1” and “BIAS2” respectively.A path for that current is through the antenna. Thus, the antenna isbiased to a high potential Vdd. The array of capacitors in FIGS.8.0101AA-CB is a series capacitance that couples the output of theSchottky detector to the input of the video amp “videoamp1” and allowsan independent bias level to be set at the input of the video amp“videoamp1.” The value of that capacitor in conjunction with theeffective resistance seen looking into the amplifier “videoamp1”determines the high pass response of the amplifier “videoamp1.” Thevalues of the capacitor and effective resistance determine the lowestfrequency at which the amplifiers can respond, and that frequency isselected to be low enough so that none of the information contained inthe base band signal is lost.

FIGS. 8.0102AA-BD provide a circuit drawing of the CMOS detector“cmosdet” which is employed in an alternative embodiment.

Details of Ouick Bias AC-Coupled Video Amplifier

FIG. 48 provides a simplified circuit schematic of a quick biasAC-coupled video amplifier 270. The video amplifier goes from a powerdown (unbiased) state to a fully biased state quickly despite a largevalue effective resistance and capacitor used to bias and couple theamplifier.

The video amplifier 270 has an input adapted to be connected to V_(in)and includes coupling capacitors 292 and 294 at the input.

The video amplifier includes a voltage divider 276 including tworesistors 278 and 280 in series, and four transistors 282, 284, 286, and288 shown to the right of a voltage divider in FIG. 48. Transistors 286and 288, the rightmost two of the four transistors, are long L (length),narrow W (width) p-channel devices operated in linear mode to providevery high effective resistance R_(EFF). Transistors 286 and 288 are usedinstead of resistors because it is hard to provide high resistancesusing resistors without generating undesirable parasitic capacitance andwithout taking up more space on an integrated circuit die. The videoamplifier 270 includes a differential amplifier 290. The voltage divider276 sets a bias voltage at the inputs of the differential amplifier 290.The effective resistance R_(EFF), in conjunction with the value ofcoupling capacitor 292 or 294, sets the angular high pass roll offfrequency for the amplifier according to a relationship ofω_(HP)=1/((R_(EFF)+R1∥R2)C1) where ω is angular frequency (2π timesfrequency), R1 and R2 are the values of the resistors 278 and 280included in the voltage divider 276, and C1 is the value of one of thecoupling capacitors. The values of R_(EFF), and the coupling capacitorsare adjusted to achieve the desired high pass roll off frequency ω_(HP)as illustrated in FIG. 49. The high pass roll off frequency determineswhat frequencies will be amplified or attenuated. The high pass roll offfrequency is set low enough so that important data is not excluded.

In many applications, the values of these components are high. Forexample, in the integrated circuit 16, R_(EFF) is approximately twoMegaOhms, and the capacitance of each of the coupling capacitors 292 and294 is approximately one picoFarad, which gives an angular high 8 passfrequency of approximately 1/((2 MegaOhms)(1 pF))=500kiloradians/second, or a high pass frequency of 500/2π=79.6 kHz.

In a powered down state, input V_(reg) is zero. Upon power up, there isa delay before the inputs reach the desired bias voltage, according to arelationship V_(BIAS)=R1/(R1+R2)V_(reg). The time constant equalsR_(EFF)C1 which is approximately equal to two microseconds.

If it is decided to wait five time constants, this requires about tenmicroseconds.

In accordance with the invention, transistors 282 and 284 are added (thetwo leftmost transistors of the four). These are short L (length) wide W(width) devices which allow the bias voltage to be established in muchless time by shorting around the high resistance of the right twotransistors 286 and 288. The time constant is thereby reduced. Thisshorting occurs when an input RXEN is low. Before using the circuit asan amplifier, RXEN is taken high (after bias. voltage is achieved). Thisrestores the desired frequency behavior.

FIGS. 8.0103AA-CF provide a circuit drawing of the video amp“videoamp1.” The video amp “videoamp1” is a differential amplifier witha cascode device isolating a resistor load from differential transistorsof the amplifier. This lowers capacitance and improves the frequencyresponse of the amplifier. Bias is provided by a resistor divider shownon the upper left of FIGS. 8.0103AA-CF, which resistor divider providesa potential to two p-channel transistors found almost in the center ofthe FIGS. 8.0103AA-CF. Each of these p-channel transistors defines avery large resistance, effectively on the order of one to two megaOhmsconnecting to the nodes of the amplifier to provide the bias. Theremaining p-channel devices shown left of center in FIGS. 8.0103AA-CFare shorting devices which short out the two p-channel transistorsduring the period when the receiver is being powered on out of the sleepmode. The function of these remaining p-channel devices is to cause theinputs to the receiver to come up to the bias level as quickly aspossible. They are then shut off in order to leave the circuit with onlythe high resistance p-channel devices providing the bias. This isnecessary from a frequency response standpoint.

FIGS. 8.0104AA-BC provide a circuit drawing of the video amp“videoamp2.” The video amp “videoamp2” operates in a manner similar tooperation of the video amp “videoamp1.” The video amp “videoamp1” has ahigher bias current than the video amp “videoamp2.” The reason for thisis to minimize the noise generated in the amplifier.

FIGS. 8.0105AA-EE provide a circuit drawing of the comparator“comparator.” The comparator has biasing considerations similar to thebiasing considerations for the video amps, and has a biasing networkshown at the left in FIGS. 8.0105AA-EE, at the inputs, which is similarto the biasing networks in the video amps. The function of thecomparator “comparator” is to ensure an output at a full digital level.

FIGS. 8.0106AA-CD provide a circuit drawing of an RF detect circuit“rxdet.” This circuit generates an RF detect signal. The circuitincludes an input switch that is either high or low, and a capacitor. Ifthe input switch is high for a sufficient percentage of the time, theinput switch will charge up the capacitor. The capacitor has a continualdischarge leakage current. As long as the input switch is highsufficiently frequently, the input switch will overcome the capacitorleakage current, and the circuit “rxdet” will put out an RF detectsignal.

FIGS. 8.0107AA-GN provide a circuit drawing showing construction detailsof the receiver bias generator “rxbias.” The receiver bias generatorincludes a series of current mirrors to produce bias currents forvarious stages of the receiver.

FIGS. 8.0108AA-AC provide a circuit drawing showing construction detailsof a data transition detector “datatx.” The data transition detector hasan input connected to the digital level output of the comparator“comparator” of the receiver “rx” in FIGS. 8.01AA-DE. The datatransition detector generates a high going pulse every time there is atransition from high to low or from low to high in the data output bythe comparator “comparator.” These transitions are tested by othercircuitry, described below, to determine whether or not a valid signalis being received.

Details of Low Power Freguency Locked Loop

As previously discussed, the integrated circuit 16 periodically checksif a radio frequency signal is being received by the receiver. Theintegrated circuit 16 includes a timer setting the period for thechecking, the timer comprising a frequency locked loop “lpfll.”

The low power frequency locked loop “lpfll” is shown in greater detailin FIGS. 8.02AA-BC. This is also shown in a simplified schematic in FIG.24. In the embodiment shown in FIG. 24, the device 12 includes afrequency locked loop (or phase locked loop) 54, a divider 55 coupled tothe input of the loop 54, and a divider 56 coupled to the output of theloop 54. A clock (e.g. 9.54 MHz) that is recovered from an incomingradio frequency command from the interrogator 26 is supplied to thefrequency locked loop (or phase locked loop) 54 after being passedthrough the divider 55. The terms “phase locked loop” or “frequencylocked loop” as used herein are meant to describe physical structure,not a state of operation. The term “locked” does not imply that thecircuitry is operating, or functioning in a locked condition. Thus, asused herein, “locked” is a term for assisting definition of a particularcircuit configuration and is not meant to imply a required state ofoperation for the circuit. To avoid ambiguity, the appended claims usethe terms “phase lock loop” or “frequency lock loop” instead of “phaselocked loop” or “frequency locked loop” to indicate that state ofoperation is not being claimed.

Phase locked loops and frequency locked loops are similar to oneanother, except that a phase locked loop tracks phase as well asfrequency. A phase locked loop includes a phase detector having a firstinput receiving the incoming message, having a second input, and havingan output; a loop filter having an input coupled to the output of thephase detector and having an output; a voltage controlled oscillatorhaving an input coupled to the output of the loop filter, and having anoutput defining an output of the phase locked loop; and a divider havingan input coupled to the output of the voltage controlled oscillator andhaving an output connected to the second input of the phase detector.The phase detector produces an output voltage proportional to the phasedifference of two input signals. The loop filter is used to control thedynamics of the phase locked loop. The voltage controlled oscillatorproduces an AC output having a frequency proportional to input controlvoltage. The divider produces an output signal having a frequency thatis an integer division of the input signal. The loop filter includes acapacitor on a control node of the voltage controlled oscillator.

The frequency locked loop 54 includes a frequency comparator 57receiving the divided recovered clock, an up/down counter 58 connectedto the output of the frequency comparator, and a current controlledoscillator 59 connected to the output of the up/down counter. The outputof the current controlled oscillator 59 is fed back to the frequencycomparator 57, and to the divider 56. The divider 56 is programmable (inresponse to a radio frequency command from the interrogator 26) in oneembodiment of the invention. To conserve power, the loop 54 is enabledonly during processing of a command from the interrogator 26, duringwhich time a recovered clock reference signal is available. In theillustrated embodiment, the current controlled oscillator 59 is a lowpower current controlled oscillator “lpcco” shown in FIGS. 8.0204AA-EJ.

FIGS. 8.02AA-BC provide a circuit drawing of the low power frequencylocked loop “lpfll.” This circuit generates a clock which is used inmultiple places to time the interval between wake ups. The clock is usedas a reference for the timed lock out function, and for the alarm timerwake up function. In a preferred embodiment, the low power frequencylocked loop “lpfll” generates a 8 kHz clock. The low power frequencylocked loop includes a current controlled oscillator “lpcco” thatconsumes very little current and that runs continuously from the timepower is first supplied to the integrated circuit 16 (“power up”) untilpower is removed from the integrated circuit 16. During power up, thelow power frequency locked loop “lpfll” attempts to synchronize to themain clock recovery oscillator “dcr” (described below). However, thatoscillator is not calibrated to anything yet because it has just beenpowered on. Still, an initial frequency is set for the low powerfrequency locked loop “lpfll.” On the first successful communicationwith an interrogator, the low power frequency locked loop “lpfll” isactually calibrated to a known clock frequency and set to a desiredfrequency (8 kHz in the illustrated embodiment).

The low power frequency locked loop includes a divider shown at the topof FIGS. 8.02AA-CB. The divider divides down an input clock signal. Inthe illustrated embodiment, the input clock signal is a 9.5 MHz clocksignal. The input clock signal is divided down by a desired factor toget a reference clock for the actual loop shown at the bottom of FIGS.8.02AA-CB. In the preferred embodiment, the reference clock for the loopshown in FIGS. 8.02AA-CB runs at 8 kHz. The loop receives a loop enablesignal “LoopEN”, shown at the lower left of FIGS. 8.02AA-CB. The loopenable signal “LoopEN” enables this frequency locked loop to operate ina loop configuration. The loop enable signal “LoopEN” is asserted when avalid message has been certified and on power up. Those are the only twotimes the loop enable signal is asserted.

The frequency of the current controlled oscillator “lpcco” is determinedby current input into the current controlled oscillator “lpcco.” Aselection of the number of current steps for controlling the oscillatoris made by the outputs of the up/down counter “udcounter.” The up/downcounter has outputs select 1 “Sel1,” select 2 “Sel2,” select. 4 “Sel4,”and select 8 “Sel8.” The outputs of the up/down counter are labelledaccording to their binary weights, and that is also how currents arerated within the current controlled oscillator. When the loop isdisabled, at the end of processing of a valid command, the count on theoutput of the up/down counter is frozen so that the select 1 throughselect 8 lines remain constant and they keep that same frequency in thelow power frequency locked loop “lpfll” until the next valid command isprocessed. At the time when the next valid command is processed, if theclock frequency of the low power frequency locked loop has drifted, theloop sets the frequency back to the desired frequency (e.g., 8 kHz).

FIGS. 8.0201AA-AB provide a circuit drawing showing construction detailsof a timed lockout divider cell “tldcel_bypass” included in the circuitof FIGS. 8.02AA-BC.

FIGS. 8.0202AA-CD provide a circuit drawing of a frequency comparator“freqcomp” of the frequency locked loop “lpfll.” The frequencycomparator counts a certain number of cycles of the reference clock andalso counts how many cycles of the low power clock occurred within thatnumber of cycles. The frequency comparator thus determines whether thelow power clock “lpfll” is running too fast, too slow, or on time. Ifthe clock is running too fast or too slow, the frequency comparatormakes an adjustment by causing the counter “udcounter” to either countup or count down. If no adjustment is necessary, the frequencycomparator makes no adjustment to the counter.

FIGS. 8.0203AA-BC provide a circuit drawing showing construction detailsof the up/down counter “udcounter” included in the low power frequencylocked loop “lpfll.” The counter has some logic on the counter's outputso that if the counter counts all the way down to zero, the counter doesnot wrap around and go to all ones. Instead, the counter stops at zero(until a signal requesting an up count is received). Similarly, if thecounter counts all the way up to all ones, the counter does not wraparound to all zeros. Instead, the counter stops at all ones (until asignal requesting a down count is received).

FIGS. 8.020301AA-BB provide a circuit drawing showing constructiondetails of an adder “udcounter adder” included in the up/down counter.

FIGS. 8.020302AA-AB provide a circuit drawing showing constructiondetails of a D type flip-flop “udcounter dff” included in the up/downcounter.

Details of Low Power Current Controlled Oscillator

The integrated circuit 16 includes the low power current controlledoscillator “lpcco.” The oscillator consumes very little current (e.g.,less than 100 nA). The oscillator “lpcco” includes digital input lines,and oscillates at a frequency controlled by the digital input lines. Thecircuit includes a thermal generator, a digitally controlled currentmirror, an oscillator, and an output driver.

FIGS. 8.0204AA-EJ provide a circuit drawing of the low power currentcontrolled oscillator “lpcco.” The low power current controlledoscillator “lpcco” includes a thermal voltage generator, including astring of resistors, shown in FIGS. 8.0204AA-EJ in the upper leftcorner. The thermal voltage generator generates a small voltageproportional to kT/q across the string of resistors where k isBoltzmann's constant, 1.38×10⁻²³ Joules per degree Kelvin, T istemperature in degrees Kelvin, and q is the electron charge in Coulombs.The voltage kT/q is approximately equal to 26 mV at room temperature.That small voltage divided by the resistor value sets the current in thecircuit. This current is approximately equal to (kT/qR) In((W/L)Q1)/(WIL)Q2). In the illustrated embodiment, the current is set toa low value (e.g., approximately three nano-amps).

Thermal generators are known in the art. See for example, “CMOS AnalogIntegrated Circuits Based on Weak Inversion Operation” by Eric Vittozand Jean Fellrath, IEEE Journal of Solid-State Circuits, Vol. SC-12, No.3, June 1977. See particularly FIG. 8 of this article, and theassociated description.

The low power current controlled oscillator “lpcco” also includes a wakeup circuit shown to the far left of the thermal voltage generator thatcauses a much higher current to flow initially to turn on the feedbackloop. The wake up circuit then shuts off and leaves the low value(nano-amp) current flowing. Thus, initialization occurs on power up andthe wake up circuit is off after that unless power is removed andreapplied. The outputs of the up/down counter “udcounter,” select 1“Sel1,” select 2 “Sel2,” select 4 “Sel4,” and select 8 “Sel8” come intothe low power current controlled oscillator as shown on the left edge ofFIGS. 8.0204AA-EJ. The low power current controlled oscillator furtherincludes control circuitry shown on the bottom strip of FIGS.8.0204AA-EJ. These outputs of the up/down counter control the number ofcurrents that are mirrored into this control circuitry by a currentmirror. The current mirror is digitally controlled and weightings arebinary in the illustrated embodiment; however, any weighting scheme canbe used. The current mirror includes transistors operating in thesubthreshold, or weak inversion mode, due to the extremely low currentlevel.

More particularly, referring to FIGS. 8.0204AA-EJ, there are fivetransistors to the right of the string of resistors, mirrored down toone about the center of the page providing a divide by five. Current isthen mirrored through all the p-channel devices. The block shown in theupper right of FIGS. 8.0204AA-EJ is a selectable current mirror. Thefirst stage generates one current equal to the reference current andthat is always flowing into the n-channel diode down at the bottom ofthat stack. Shown to the right of the n-channel diode are the selectablegroups of p-channels. The first one has one, the next two, the nextfour, the next eight in a binary sequence. The transistors shown belowthe p-channels transistors are select devices and they are controlled bythe digital signals select 1 “Sel1,” select 2 “Sel2,” select 4 “Sel4,”and select 8 “Sel8.” Thus, the number of currents can be selected andhowever many are selected are added into the one that is always flowingin the diode.

The current from groups of p-channel transistors that are not selectedis diverted over to a separate or second diode shown on the far right ofFIGS. 8.0204AA-EJ. This is so that when a block of transistors is notselected, their drain nodes do not get pulled up to VDD. By sinking thecurrent in this second diode, the voltage at the drain node of anunselected block of transistors is kept down near the voltage at whichit will operate when and if it is actually connected over to the firstdiode. This is so that, upon switching a select line, a capacitancedoesn't have to be charged from V_(DD) down to the proper operatingvoltage. In any case, the selected number of currents are added togetherinto the first diode, and then that voltage is carried on the line shownin FIGS.8.0204AA-EJ as going down the right side of page, which line hasa capacitor tied to it. That capacitor is a filter capacitor so that thevoltage on that node does not change abruptly when the select lineschange or when some unrelated signal nearby switches. Since all of thesetransistors are operating in a sub-threshold or weak inversion mode, asmall change in the voltage on their gate will otherwise result in arather large impact on the operation of the circuit. That line is theinput for the circuitry shown across the bottom of FIGS.8.0204AA-EJ.There is a current mirror situation there, SO that the sum of all theselected currents plus the one default current flowing in the diodeabove is mirrored and flows through the p-channel devices of thiscircuitry. There is then another mirror to generate bias voltages forthe n-channel current source devices for the VCO ring oscillator. Thep-channel gate voltages are used in mirroring into the p-channel loaddevices of the same ring oscillator. The frequency of this ringoscillator is controlled by the current mirrored to them.

The low power current controlled oscillator includes a four stage ringoscillator. The frequency of oscillation is approximately proportionalto the amount of current flowing. The frequency of oscillation of thefour stage ring oscillator is directly proportional to its bias currentover a wide range of frequencies. For example, frequency is directlyproportional to bias current for frequencies between approximately 100Hz and tens of MHZ (e.g. to twenty MHZ).

The low power current controlled oscillator further includes an outputdriver. In the illustrated embodiment, the output driver includes acomparator circuit receiving the output of the fourth stage ring. Thepurpose for this comparator is to convert the small output signal of theoscillator to full digital levels. In the illustrated embodiment, fulldigital levels are zero volts and V_(DD). In the illustrated embodiment,V_(DD) is 3.3 Volts ±0.3 Volts. In an alternative embodiment, V_(DD) is5 Volts ±10 or 20%. Another other suitable values can be employed forV_(DD) and the digital levels.

Circuitry is included to eliminate the crossover current in the n and pchannel devices in the first few invertors. This is because, whenoperated at very low current levels, the rise and fall times are longand could allow substantial current to flow in the n and p channeldevices during switching.

The digital levels are buffered and amplified up by the comparator toprovide an output signal from the low power frequency locked loop“lpfll.” The output of the low power current controlled oscillator isshown on the right edge of FIGS. 8.0204AA-EJ. In the illustratedembodiment, the low power current controlled oscillator operates ateight kHz. However, if desired for alternative embodiments, the lowpower current controlled oscillator is capable of running at a frequencyfrom approximately 100 Hz to 20 kHz. In an alternative embodiment, thelow power current controlled oscillator is capable of running at afrequency from approximately 100 Hz to 30 kHz.

The low power current controlled oscillator consumes very little power.For example, in the illustrated embodiment, the low power currentcontrolled oscillator consumes less than a milliAmp. More particularly,in the illustrated embodiment, the low power current controlledoscillator consumes approximately 100 nanoAmps.

In an alternative embodiment, instead of using a thermal voltagegenerator, a transistor is biased in the subthreshold region in order todefine the current source and to generate a small current. However, inthis embodiment, the voltage on the gate of the transistor is updatedperiodically as it leaks away.

Although the low power current controlled oscillator has been describedin connection with a radio frequency identification device, the lowpower current controlled oscillator can be advantageously employed inany battery powered electronic product which must keep track in time.

FIGS. 8.03AA-AB provide a circuit drawing showing construction detailsof a counter bit “lpfll_cbit” included in the receiver “rx.”

FIGS. 8.04AA-EE provide a circuit drawing of the wake up controller“rxwu.” An input to the wake up controller is a clock signal “LPCLK”from the low power frequency locked loop “lpfll.” This clock signalinput is shown in the upper left of FIGS. 8.04AA-EE. The clock signal“LPCLK” is further divided down to provide certain time intervalsavailable for selection. These are the time intervals at which theintegrated circuit 16 will wake up and look for a radio frequencysignal. In the illustrated embodiment, these time intervals are set at0.5, 16, 64 and 256 milliseconds. The selection of one of these multipleavailable time intervals is accomplished via radio frequency commandfrom the interrogator.

The wake up controller includes wake up abort logic shown in the lowerleft of FIGS. 8.04AA-EE. The wake up abort logic performs a number oftests (described elsewhere herein) to determine whether the receivedsignal is a valid signal and, if all tests are passed, then the wake upcontroller asserts a signal on line “RXWU” shown on the right of FIGS.8.04AA-EE. This signal wakes up the processor, and the processor thenprocesses the command contained in the message.

Details of Wake Up Tests

FIGS. 8.0401AA-AB provide a circuit drawing of a wake up abort logiccircuit “wuabort.” The wake up abort logic circuit provides forconservation of battery power. If what is received is not a validmessage, the wake up abort logic circuit determines this quickly and thedevice returns to the sleep mode so that the battery is not drained oninvalid messages or spurious communication. The wake up abort logiccircuit works by counting clock cycles. The wake up abort logic circuithas as an input a clock signal “CHIPCLK” that is the output of the clockrecovery oscillator “dcr” (described below). This clock signal isdivided down by a factor of four, which results in a value approximatelyequal to the spread spectrum chip rate. After the clock is actuallyacquired from a message from the interrogator, the resulting value willbe equal to the chip rate.

Initially though, when these wake up tests are performed, a clock hasnot yet been acquired from a message. The wake up abort logic includesan RF Detect Timer, shown on the top, left of FIGS. 8.0401AA-EE, whichperforms a first test. The RF Detect Timer counts a predetermined numberof cycles of the clock (e.g., 13 cycles) and, if the RF detect signalfrom the receiver is not asserted, the wake up is aborted. On the otherhand, if the RF detect signal is asserted within those cycles, the wakeup abort logic starts the next series of tests without waiting for theend of the predetermined number of cycles.

The next series of tests are timed by a counter shown across the centerof the page in FIGS. 8.0401AA-EE. For the next tests, transitions in theincoming data stream are counted within a certain time interval and thenumber of transitions must fall within a certain range in order to passthe test. Transitions are counted by the counter shown at the bottom inFIGS. 8.0401AA-EE. The range limits are set by knowing the number oftransitions that should occur in the data within the amount of timeallowed. This is known because each data bit is encoded as a thirty-onechip sequence as described elsewhere herein. The reason there is a rangeis because the clock has not yet been acquired accurately so there is arange of clock frequencies that must be considered. In the illustratedembodiment, one test checks whether, after five counts of the clock,there has been between greater than or equal to one, and less than eighttransitions in the data. If not, the wake up is aborted and the devicegoes back to sleep. If yes, then the next test is performed.

The next test checks whether, after twenty-six clock counts, there aregreater than or equal to fourteen and less than thirty-two transitionsin the data. If not, wake up is aborted and the chip goes back to thesleep mode. If this test is passed, the wake up abort logic circuitperforms tests relating to signals generated by the clock recoverynodes. One such test is a test for chip lock. Chip lock is an indicationthat clock recovery is proceeding and has actually gotten within a fewpercent of the desired clock frequency. The final check is whetherfrequency lock has occurred. Again, these tests are timed. If one of thesignals is not asserted by the time the timer signal goes high, then thewake up will be aborted and the device goes back to sleep and will tryagain after another wake up interval. Frequency lock will come into thelogic in the center of the page in FIGS. 8.04AA-CB, and that is whatcauses the RXWU signal to be asserted, thus waking up the processor.

Another function of the wake up abort logic shown in FIGS. 8.0401AA-EEis to discriminate between high rate and low rate. The wake up abortlogic measures time while these tests are performed to determine whenthe interrogator is in high rate, but the chip is in low rate or viceversa and abort out of wake up (return to the sleep mode).

These tests will now be described in connection with flowchartsillustrated in FIGS. 25-27.

The wake up controller “rxwu” was described above in connection withFIGS. 8.04AA-EE. The wake up tests performed by the wake up controllerare illustrated in flow chart form in FIGS. 25-26.

When the integrated circuit 16 first wakes up, bias generators and thereceiver “rx” are powered on (step Si in FIG. 25). After ensuring thatthe bias is on (step S2 in FIG. 25), the master clock “dcr” is started.By design, the master clock “dcr” starts at a frequency below the finalfrequency it will achieve after the clock recovery circuit “lpfll”extracts the clock frequency from the incoming signal. Moreparticularly, in the illustrated embodiment, the master clock starts ata start frequency above half of the final frequency it will achieveafter the clock recovery circuit “lpfll” extracts the clock frequencyfrom the incoming signal. Still more particularly, in the illustratedembodiment, the master clock starts at a start frequency between halfand three quarters of the final frequency the master clock will achieveafter the clock recovery circuit “lpfll” extracts the clock frequencyfrom the incoming signal. In the illustrated embodiment, the finalfrequency is 38.15 MHZ, and the start frequency is between 20 and 30MHZ. The master clock includes a frequency locked loop including avoltage controlled oscillator. An offset is applied to the oscillator tomake sure that the clock starts at least as fast as 20 MHZ. Then, thefrequency locked loop adjusts the frequency to 38.15 MHZ.

Because the clock has not yet been acquired from the incoming signal,the clock is a free running oscillator when providing the startfrequency. Initial wake up tests are performed at this lower startfrequency. The receiver “rx,” digital clock and data recovery circuit“dcr,” pseudo random number processor “pnproc,” and voltage controlledoscillator “vco” are turned on (step S3 in FIG. 25).

The input radio frequency signal received from the interrogator 26 is adirect sequence spread spectrum input signal in the illustratedembodiment. Spread spectrum techniques are described above. In oneembodiment, incoming radio frequency commands are included in packetsthat contain, in order of transmission, a preamble, a Barker code, andthe command. In one embodiment, each bit of the incoming radio frequencycommand sent by the interrogator is modulated using a pseudo noise (PN)sequence for direct sequence spread spectrum communication.

After the clock is running, the device 12 is in a receiver on modeillustrated in FIG. 27 by a vertical line marked “WAKEUP RX ON.” Afterthe clock is running, the device 12 performs wake up tests (at the loweror start frequency).

A first test is whether the receiver “rx” detects any radio frequencysignal within a predetermined number of clock cycles (step S4 in FIG.25). In the illustrated embodiment, this predetermined number of clockcycles is 13. If no radio frequency signal is detected by the receiver“rx” within 13 clock cycles, the device 12 returns to the sleep mode. Ifa radio frequency signal is detected by the receiver “rx” within 13clock cycles, the device 12 switches to a wake up abort test modeillustrated in FIG. 27 by a vertical line marked “WAKEUP ABORT TESTS,”and a second test is performed.

In the second test, a determination is made as to whether apredetermined number of data transition pulses occur within apredetermined number of clock pulses for the radio frequency signaldetected by the receiver “rx” (step S5 in FIG. 25). More particularly,the device 12 includes a long counter shown in FIGS. 8.0401AA-EE drivenby a clock signal “CHIPCLK.” The device 12 further includes a circuit“datatx” which detects transitions in the signal received by thereceiver “rx” and generates a pulse (“DTX” in FIG. 26) at eachtransition. The device 12 further includes another counter circuit shownin FIGS. 8.0401AA-EE which counts these pulses. Because a valid incomingsignal is modulated with a known PN sequence, the number of transitionsin a given time for a valid incoming signal is known. The device 12includes logic “wuabort” that tests whether the proper number of datatransition pulses occur within a certain number of clock pulses. Moreparticularly, in the illustrated embodiment, the logic tests whethermore than or equal to one and less than eight such data transitionpulses occur within five chips. If not, the device returns to the sleepmode. If so, a third test is performed.

In the third test, a determination is made as to whether a predeterminednumber of data transition pulses occur within a predetermined number ofclock pulses for the radio frequency signal detected by the receiver“rx” (step S5 in FIG. 26). The third test is similar to the second test,except that the number of data transition pulses is tested against anumber of clock pulses that is different from the number in the secondtest. More particularly, in the illustrated embodiment, the logic testswhether more than or equal to fourteen and less than thirty-two suchdata transition pulses occur within thirty-one chips. If not, the device12 returns to the sleep mode.

If the above transition tests are passed, the device 12 checks to see ifthe clock recovery circuit locks onto the incoming clock rate. Moreparticularly, in the illustrated embodiment, a determination is made asto whether a clock is acquired from the incoming signal within 6 k chips(step S7 in FIG. 26). A determination is then made as to whetherfrequency lock is achieved within 16 k chips (step S9 in FIG. 26). Thedevice 12 returns to the sleep mode if any of these tests fail. If thesetests are passed, then the device 12 enters a processor on modeillustrated in FIG. 27 by a vertical line marked “PROCESSOR ON.” Poweris supplied to the processor (step S10 in FIG. 26) and the device 12waits for the preamble of the incoming message to end and the command tobegin.

In one embodiment, the tests of FIG. 26 are employed to distinguishbetween incoming signals with different possible valid chipping rates.

More particularly, in the illustrated embodiment, it is known how longeach of the various tests should take for valid low chipping rate orhigh chipping rate signals, and this information can be tested todetermine whether the incoming signal is a high rate or low rate signal.

Other appropriate tests can be performed in embodiments where spreadspectrum is not employed. In these embodiments, knowing how valid datais encoded, the wake up timer and logic still compares the number oftransitions received in a given amount of time with an expected numberof transitions for a valid signal.

FIGS. 8.040101AA-AB provide a circuit drawing showing constructiondetails of a counter bit “wuabort-cbit” included in the wake up abortlogic.

FIGS. 8.0402AA-AB provide a circuit drawing showing construction detailsof a timed lockout divider cell “tldcel” included in the receiver wakeup controller.

Details of Lock Detection in a Digital Clock Recovery Loop

In many communications systems, it is necessary to recover a clocksignal from the received data. A phase locked loop is one way ofrecovering such a clock signal. In the illustrated embodiment, such arecovered clock is used as a master clock.

The integrated circuit 16 includes the digital clock and data recoverycircuit “dcr” which includes a phase locked loop. The phase locked loopincludes a voltage controlled oscillator “dcr_vco.” The frequency of thevoltage controlled oscillator always starts low, at between 50% and 75%of the final desired value. When the voltage controlled oscillatorstarts running, large steps are taken (FIG. 54). As the frequencyapproaches the final value, increasingly smaller steps are taken toachieve greater accuracy. The illustrated embodiment employs four stepsizes: large, medium, medium-fine, and fine. For example, in theillustrated embodiment, large steps up are employed between 50% to 75%of the final desired value, and medium steps up are then taken above 75%until pump up commands are not issued for a predetermined number oftransitions, then medium-fine steps up are employed until the finalvalue is overshot, then fine steps down are employed.

A method is needed to determine when the frequency of the voltagecontrolled oscillator matches the desired frequencies contained in thereceived data.

The voltage controlled oscillator includes a control node having avoltage indicative of the frequency of the voltage controlledoscillator. The behavior of this node is used to determine whenfrequency lock has occurred.

After the phase locked loop has run long enough to get within a fewpercent of the final value (at a time illustrated as To in FIG. 54), asignal “SDD” (start data decoding) is generated. This signal “SDD”disables the large and medium steps and enables lock detect circuitryfor determining if frequency lock has occurred. A latch “KILLSU” (killstart up) detects when the first fine step pump down occurs (at T₁ inFIG. 54). This enables a latch “FREQLOCK.” The latch “FREQLOCK” is setwhen the first fine step pump up occurs (at a time illustrated as T₂ inFIG. 54). A signal “FREQLOCK” is then indicative that the phase lockedloop has reached its final value.

In other words, large, medium, then medium-fine steps up are followed byfine steps down. The final value is overshot, and a frequency locksignal is provided upon occurrence of the first subsequent fine step up.

In the illustrated embodiment, the final value of the voltage on thecontrol node of the voltage controlled oscillator, where frequency lockis expected, is approximately 1.2 Volts. In one embodiment, each largestep is approximately several hundred milliVots, each medium step has asize approximately in the tens of milliVots (e.g., 25 milliVots), eachmedium-fine step has a size of approximately a few milliVots (e.g. twomilliVots), and each fine step has a size approximately in the tenths ofmilliVots. Various other relative sizes or numbers of steps are employedin alternative embodiments.

The sizes of steps is set using current sources of different values thatare turned on for a fixed period of time to drive to the capacitor onthe control node of the voltage controlled oscillator.

In the illustrated embodiment, the fine step generator is not disabledbefore time T₀ so there is a possibility that a combination of finesteps with larger steps can take place before time T₀. In an alternativeembodiment, however, the fine step generator is disabled before time T₀.

FIGS. 8.05AA-CB provide a circuit drawing of the digital clock and datarecovery circuit “dcr.” The digital clock and data recovery circuitincludes a phase locked loop of a digital design, and a state machine“dcr_statemachine” that drives the phase locked loop. The phase lockedloop includes a voltage controlled oscillator “dcr_vco” and controlcircuitry “dcr_vcocontrol” for the voltage controlled oscillator. Thevoltage controlled oscillator “dcr_vco” includes a control node (“OUTN”and “OUTP” shown in FIGS. 8.0504AA-EE and described below in greaterdetail) and produces an oscillation at a rate dependent on the value ofa voltage applied to the control node. In the illustrated embodiment,the state machine has four states. The phase locked loop produces anoutput pulse on a line “OUTC” (later labeled “FMASTER”). The digitalclock and data recovery circuit attempts to place four pulses of theoutput clock within one chip time.

The state machine “dcr_statemachine” determines when that is not thecase and, if not, whether to cause the oscillator to run faster or torun slower. The state machine “dcr_statemachine” then issues appropriatepump up or pump down signals to drive a control node of the oscillator.The voltage controlled oscillator “dcr_vco” starts out at a minimumfrequency as determined by an offset current which is present regardlessof the loop. This ensures that the oscillator will start up and run atgreater than 50% of the final value so that the phase locked loop willconverge on the proper frequency. The digital clock and data recoverycircuit also includes a PLL start-up circuit “dcr_startup.” Theacquisition of the clock frequency happens in stages and, initially, thecontrol node moves in large increments towards its final value. Thestart-up circuit “dcr_startup” provides large increments for controllingthe loop. However, as the digital clock and data recovery circuit getscloser to acquisition of clock frequency, control switches from thatstart up circuit “dcr_startup” over to the state machine“dcr_statemachine.” The state machine provides very fine steps as thefinal convergence is done with very fine steps. The data stream is fedinto the circuitry on the upper right. Then the data is sampled duringone of the states of the state machine after it has been determined thatthe data is valid. The data stream is recreated and called “RXCHIPS.”

FIGS. 8.0501AA-BE provide a circuit drawing of the start up circuit“dcr_startup” included in the digital clock and data recovery circuit.In the illustrated embodiment, the start up circuit provides either verylarge or fairly large steps dependent upon how far from frequency theoscillator is running. The start up circuit also has a counter (shownalong the bottom in FIGS. 8.0501AA-BE) that determines when there havebeen no pump up commands issued for a given count of transitions. In theillustrated embodiment, the counter determines when there have been nopump up commands during sixteen transitions. If the given count oftransitions are detected in the data and there has been no pump upcommand (e.g., no pump up medium or pump up fast command) then a signalis asserted on a line “SDD.” SDD stands for Start Data Decode and is anindication that the control voltage has converged to within a fewpercent.

FIGS. 8.050101AA-BE provide a circuit drawing showing constructiondetails of a shift register cell “dcr_sreg” included in the PLL start upcircuit. FIGS. 8.050102AA-AB provide a circuit drawing showingconstruction details of a counter bit “dcr_counterbit” included in thePLL start up circuit.

FIGS. 8.0502AA-CD provide a circuit drawing of the state machine“dcr_statemachine.” In the illustrated embodiment, the state machine hasfour states. The state machine includes two flip-flops with feedbacksignals providing the four states. This circuit generates pump up slow,and pump down slow commands for adjusting voltage on a control node ofthe voltage controlled oscillator.

This circuit also has the circuitry that turns off the start up circuitand generates the frequency lock signal. When trying to acquirefrequency lock, there will be large and medium pump ups, without anypump downs, until the final desired value is overshot. At this point,there will be a first pump down slow pulse. When the first pump downslow command is issued, the start up circuitry “dcr_startup” is turnedoff, which leaves only fine step capability for adjustment in thecontrol voltage. It takes time for the fine steps to bring down thecontrol node voltage to the proper value and the voltage on the controlnode will overshoot the desired voltage in the negative direction. Thestate machine will detect that it has gone too far and it will step thevoltage back up towards the final value and that first fine step up willbe detected and at that point the frequency lock signal is asserted.

FIGS. 8.0503AA-BB provide a circuit drawing of a bias generator“dcr_bias.” The bias generator includes current mirrors that generatethe appropriate bias values for the various circuits in the digitalclock and data recovery block.

The digital clock and data recove circuit “dcr” includes a VCO controlvoltage generator “dcr_vcocontrol” which is shown in greater detail inFIGS. 8.0504AA-EE.

The digital clock and data recovery circuit “dcr” employs a phase lockedloop to recover the clock frequency from an incoming radio frequencymessage. Phase locked loops use feedback to maintain an output signal ina predetermined phase relationship with a reference signal.

Details of Digital Clock Recovery Loop

Operation and design of the digital clock and data recovery circuit“dcr” will now be further described with reference to FIGS. 61-72

In many communications systems it is necessary to recover a clock signalfrom the received digital data stream. In the device 12, this clocksignal is used as the master timing reference to eliminate the need foran external crystal-based timing reference. Typically, a phase lockedloop of some type is used to extract the clock.

There are many requirements on the phase locked loop used to recover aclock signal from the received digital data stream. Several importantones for this application are that the phase locked loop must acquirethe desired frequency without locking to a multiple or submultiple ofthe desired frequency; the phase locked loop must lock to the desiredfrequency within a certain time of interest; and the phase locked loopmust yield consistent performance despite wide variation in deviceparameters which is inherent in integrated circuit processing. The phaselocked loop employed in the illustrated embodiment, erembodied in thedigital clock recovery circuitry “dcr,” satisfies all of theserequirements.

In the illustrated embodiment, the forward link baseband data is encodedfor direct sequence spread spectrum. In the illustrated embodiment, adata bit “1” is represented by a thirty-one chip sequence and a data bit“0” is represented by the logical inversion of the same thirty-one chipsequence.

The mode of operation of the device 12 is as follows. The chipperiodically awakens from a low-current sleep mode in order to detectwhether an incoming RF message is present. The clock recovery loop “dcr”is inactive in the low-current sleep mode. If a message is present, themessage is tested to make sure it is a valid message from aninterrogator. If the incoming signal passes these tests, the clockrecovery loop is enabled, the clock is acquired, the message isprocessed, and a reply is sent. The device 12 then returns to sleepmode.

The digital clock recovery loop is illustrated by reference numeral 700in FIG. 61. The digital clock recovery loop 700 comprises severalsub-circuits. The digital clock recovery loop 700 includes a voltagecontrolled oscillator 702. The voltage controlled oscillator 702 has anoutput 704, and produces a square wave at output 704 having a frequencycontrolled by the voltage on an input control node. When the voltage onthe control node is zero, the frequency at output 704 is at least onehalf of the final recovered frequency and not greater than the finalrecovered frequency. The output frequency rises monotonically, nearlylinearly, as the control node voltage is increased. This is shown inFIG. 62. More particularly, FIG. 62 illustrates the frequency producedat the output 704 of the voltage controlled oscillator 702 relative to avoltage at the input control node.

The digital clock recovery loop 700 further includes a charge pump andloop filters which control the rate of change of the voltage on thecontrol node of the voltage controlled oscillator. The charge pump andloop filters are designated in FIG. 61 with reference numeral 706.

The digital clock recovery loop 700 further includes a start-up circuit708 which performs frequency detection when the voltage controlledoscillator first starts up and, in conjunction with the charge pump andloop filters 706, causes the voltage on the control node of the voltagecontrolled oscillator to change rapidly.

The digital clock recovery loop 700 further includes a state machine 710which performs phase detection when the frequency of the voltagecontrolled oscillator is within a few percent of its final value and, inconjunction with the charge pump and loop filters, causes the voltage onthe control node of the voltage controlled oscillator 702 to changeslowly.

The only analog blocks are the voltage controlled oscillator 702 and thecharge pump. The rest of the circuits of the digital clock recovery loopare digital circuits which are easy to build at high yield in integratedcircuit processes.

In the preferred embodiment, the voltage controlled oscillator 702 isthe voltage controlled oscillator “dcr_vco” shown in the detailedschematic drawings, and has control nodes “OUTN” and “OUTP”; the statemachine 710 is the state machine “dcr_statemachine” shown in thedetailed schematic drawings; and the start-up circuit 708 is thestart-up ocircuit “dcr_startup” shown in the detailed schematicdrawings.

The digital clock recovery loop causes the frequency at the output ofthe voltage controlled oscillator to vary until a predetermined numberof this clock fit within the time interval of an identifiable discretesegment of the incoming data. More particularly, in the illustratedembodiment, the digital clock recovery loop causes the frequency at theoutput of the voltage controlled oscillator to increase until exactlyfour cycles of the clock fit within the time interval of a single chip.In alternative embodiments, other integer numbers could be used. In theillustrated embodiment, a state machine having four states is employedto cause the frequency at the output of the voltage controlledoscillator to increase until exactly four cycles of the clock fit withinthe time interval of a single chip. A general description of thebehavior of the control node voltage can be found above in the sectiontitled Details of Lock Detection in a Digital Clock Recovery Loop.

What follows is a discussion of the operation of each block of thedigital clock recovery loop. The start-up circuit 708 is show in FIG.61. Although it may be simplified from the circuitry shown in thedetailed schematics including “dcr_startup” shown in FIGS. 8.0501AA-BE,the theory of operation is the same.

The start-up circuit 708 includes a plurality of flip-flops 712 chainedtogether, a plurality of flip-flops 714 chained together, and anexclusive-or gate 716. The exclusive-or gate 716 has an output connectedto the input of the first of the flip-flops 714, has an input connectedto the output of the last of the flip-flops 712, and has another inputconnected to the input of the same flip-flop 712. More particularly, inthe illustrated embodiment, each flip-flop 712 and 714 is a D-typeflip-flop and has a D input, a clock input, and a Q output. The D inputof flip-flops 712 other than the first flip-flop is connected to the Qoutput of a previous flip-flop 712. The first flip-flop 712 is connectedto the input data “Data In.” The D input of flip-flops 714 other thanthe first flip-flop 714 is connected to the Q output of a previousflip-flop 714. The first flip-flop 714 is connected to the output of theexclusive-or gate 716. The clock inputs of the flip-flops 712 and 714are all tied to the output 704 of the voltage controlled oscillator 702.Data is shifted from the D input of each flip-flop to the Q output ofthe same flip-flop on each clock pulse. Thus, the flip-flops 712 as agroup define a shift register, and the flip-flops 714 as a group definea shift register.

The start-up circuit 708 further includes an AND gate 718 that has oneinput that is the output of the exclusive-or gate 716, has a secondinput that is the output of the second of the flip-flops 714, anddefines an output “Puf1” (a first pump up fast output). The start-upcircuit 708 further includes an AND gate 720 that has one input that isthe output of the exclusive-or gate 716, has a second input that is theoutput of the third of the flip-flops 714, and defines an output “Puf2”(a second pump up fast output).

The start-up circuit 708 further includes a counter 722 that receives asinputs “Puf1” and “Puf2” and generates an output “SDD” (start datadecode) when the output of the voltage controlled oscillator 702 isclose to its final value.

The exclusive-or gate 716 in the center of the page generates a highoutput whenever there is a transition in the data as sampled by theclock signal output by the voltage controlled oscillator 702 outputclock. Assume for discussion that data is latched into all flip-flops712 and 714 on the falling edge of the clock. Puf2 goes high when threefalling edges of the clock occur within one chip because the inputs ofthe AND gate are spaced apart by three flip-flops. Three falling edgesof the clock occur within one chip when the frequency is between 75% and100% of the final value. Puf1 goes high when two falling edges of theclock occur within one chip because the inputs of the AND gate arespaced apart by two flip-flops. Two falling edges of the clock occurwithin one chip when the clock frequency is 50% to 75% of its finalvalue. This is shown on the waveform diagram of FIG. 63 for the casewhen the frequency is exactly 50%. Puf1 could be used to pump up thecontrol node of the voltage controlled oscillator 702 rapidly. Puf2could be used to pump up the control node of the voltage controlledoscillator 702 at a rate equal to that for Puf1 (as is shown in FIG. 61)or it could pump at a slower rate (as is done in the circuitry shown inthe detailed schematics). As the clock frequency approaches 75% of finalin the Puf1 case or 100% of final in the Puf2 case, pump up signalsoccur infrequently as error must accumulate over a long time to causethe appropriate number of clock edges to shift within a chip. This isused to detect when the clock frequency is close to its final value.

The counter 722 counts transition pulses until it is cleared by a Puf1or Puf2 signal. If a predetermined large number of transitions arecounted before a pump up occurs, a signal is asserted on a line SDD(start data decode). In the illustrated embodiment, if sixteentransitions are counted before a pump up occurs, a signal is asserted online SDD. This indicates that the voltage on the control node of thevoltage controlled oscillator is within a few percent of its finalvalue, allowing data to be accurately recovered.

In the illustrated embodiment, the state machine 710 issues finerpump-up signals than the start-up circuit 708, and can also issuepump-down signals. In the illustrated embodiment, the start-up circuit708 only issues pump up signals. The state machine 710 has as manystates as the number of clock cycles which fit within one chip time. Inthe illustrated embodiment, the state machine has four states. The statemachine 710 counts clock pulses and expects the data to transition at acount of one every time there is a transition. If the transitionactually occurs at a count of four then the clock is too slow and a pumpup is issued. If the transition actually occurs at a count of two thenthe clock is too fast and a pump down is issued. If the transitionactually occurs at a count of three, it is not known whether the clockis fast or slow so no adjustment is made to the voltage to controlledoscillator. A state diagram is shown in FIG. 64.

Design of a clocked sequential circuit is known in the art. See, forexample, chapter 6 of Digital Logic and Computer Design by M. MorrisMano, 1979, Prentice-Hall, Inc. A typical design procedure involvesdescribing circuit behavior using a state diagram (see FIG. 64),obtaining a state table (see FIG. 66), assigning binary values to eachstate (see FIG. 64), determining the number of flip-flops needed (seeFIG. 65), choosing the type of flip-flops to be used (see FIG. 65),using Karnaugh maps or other simplification methods, deriving circuitoutput functions and flip-flop input functions (see FIGS. 67 and 68),and drawing the logic diagram. The numbers in parentheses in FIG. 64 arethe binary state numbers. ENDT enables the sampling of the data (alwaysat state two when no transition occurred). There are several ways toimplement a circuit to perform functions of a state diagram. Assume thatQ1 and Q0 are the binary state numbers in parentheses above (Q1 on theleft, Q0 on the right), and that D1 and D0 are the next state values ofQ1 and Q0, respectively. This is illustrated in FIG. 65. The flip-flopoutputs Q0 and Q1 are the states. Then, a state table can be derived.This is shown in FIG. 66. Using Karnaugh maps (see FIGS. 67 and 68),minimum logic to perform the desired function can be derived. It shouldbe noted, of course, that minimum logic need not be employed—logicinvolving an increased number of logic gates but performing the samedesired function can also be employed. From the Karnaugh map shown inFIG. 67, the following equation can be derived:

D0=Ql+TX·Q0+En·TX

which can also be written as:

D0=[Q1′·(TX·Q0)′·(En·TX)′]′

where the symbol “+” represents a logical OR, the symbol “·” representsa logical AND, and the symbol “′” represents a logical NOT.

From the Karnaugh map shown in FIG. 68, the following equation can bederived:

D1=TX′·Q1·Q0′+En·TX′Q0′

which can also be written as:

D1=[(TX′·Q1·Q0′)′·(En·TX′·Q0′)′]′

where the symbol “+” represents a logical OR, the symbol “·” representsa logical AND, and the symbol “′” represents a logical NOT.

Logic to implement these equations is shown in FIGS. 69 and 70.

Paths shown in FIG. 64 are defined as follows:

ENDT=Q1′·Q0·TX′

PumpUpSlow=Q1·Q0′·TX; and

PumpDownSlow=Q1′·Q0·TX

Logic used to implement the state machine, in accordance with oneembodiment of the invention, is shown in FIGS. 8.0502AA-CD.

A simplified timing diagram showing operation of the state machine isshown in FIG. 71. The crowding and separation of states in FIG. 71 isexaggerated to show the various modes of operation in a compact form.More particularly, it is highly unlikely that a pump down signal wouldbe necessary so soon after a pump up signal as is depicted in FIG. 71.

The state machine is trying to fit four cycles of the output of thevoltage controlled oscillator in one chip width. Referringsimultaneously to FIGS. 71 and 64, starting at the first occurrence ofstate 3 in FIG. 71, there is no transition, so the state machine willproceed to state 4 on the next clock. At state 4, there is notransition, so the state machine will proceed to state 1 at the nextclock. At state 1, there is a transition in the waveform. The statemachine always proceeds to state 2 from state 1. At state 2, there is notransition. From state 2, the state machine proceeds to state 3. Thiscycle is repeated and these paths are followed unless the clock recoveryloop drifts off frequency.

If the clock recovery loop drifts off frequency, other paths of thestate diagram of FIG. 64 are followed. For example, if a transition isseen at state 4, the voltage controlled oscillator is oscillating tooslowly, and a PumpUpSlow is issued. The state machine skips state 1 andgoes to state 2.

If, after going from state 1 to state 2, a transition is seen, thevoltage controlled oscillator is oscillating too fast. The state machinewill go from state 2 to state 2 so that state 2 is now in the properposition.

If a transition is seen at state 3, the voltage controlled oscillatormay either be oscillating too fast or too slowly, so no pump up or pumpdown signals are issued. Instead, the state machine proceeds to state 2.

The control functions performed by the start-up circuit and statemachines can be used to control the frequency of any voltage controlledoscillator. The particular voltage controlled oscillator 702 that isemployed in the illustrated embodiment is shown in FIGS. 8.0505AA-EF.

In the illustrated embodiment, the voltage controlled oscillator 702includes a current controlled four-stage ring oscillator shown in thecenter of FIGS. 8.0505AA-EF. The frequency of oscillation is very muchlinearly proportional to the bias current flowing in each stage.

The voltage controlled oscillator 702 further includes an OperationalTransconductance Amplifier shown on the left side of FIGS. 8.0505AA-EF.This Operational Transconductance Amplifier converts a voltagedifference at its inputs to a current difference at its outputs. ThisOperational Transconductance Amplifier has a characteristic that islinear over a range of input voltage.

The composite circuit is a voltage controlled oscillator 702 with nearlylinear operation about the operating point of 38.15 MHz. The circuitshown to the right in FIGS. 8.0505AA-EF converts the small signal outputof the oscillator to full digital levels.

The input reference voltage is generated by a bandgap regulator and hasa value of about 1.2 volts. The circuit is designed so that at nominalconditions the control node needs to pump to about equal to thereference voltage to oscillate at 38.15 MHz.

The start-up circuit requires that the oscillator start at greater thanhalf frequency (approximately 19 MHz) and less than full frequency overall operating conditions and for all process variations. This oscillatorstart frequency is set by providing an offset current to the bias of theoscillator which is not controlled by the input voltage. In theillustrated embodiment, the range of allowed offset currents is 7.437 μAto 9.763 μA. A value of 8.2 μA was chosen. Thus, the oscillator startfrequency will vary from about 20 MHz to 34 MHz.

The charge pump and loop filters 706 are shown in greater detail inFIGS. 8.0504AA-EE. The filter capacitors are shown on the right side ofFIGS. 8.0504AA-EE. In the illustrated embodiment, the filter capacitorsinclude a first group of ten capacitors, defining a total capacitance of10 pF, and an second group of ten capacitors, defining a totalcapacitance of 10 pF. In FIGS. 8.0504AA-EE, the first group of tencapacitors is shown above the second group of ten capacitors. Othervalues or numbers are possible. In the illustrated embodiment, the lowergroup of capacitors is connected to the reference voltage input to thevoltage controlled oscillator 702. The upper group of capacitors isconnected to the control node input of the voltage controlled oscillator702. The control node always starts at 0 Volts and is pumped up. Theother (reference) side is always at the bandgap voltage.

The charge pump is shown in the center of FIGS. 8.0504AA-EE. In theillustrated embodiment, there are actually four charge pumps. The methodemployed is to steer a current to charge or discharge the 10 pFcapacitor for a prescribed period of time (one cycle of the recoveredclock, in the illustrated embodiment). The change in control voltage fora single pump is:

ΔV=(I/C)Δt

The lower three of the illustrated charge pumps are controlled by thestart-up circuit 708 and can only pump up. The upper pump is controlledby the state machine 710 and can pump up or down in fine steps. The stepsizes are controlled by the current value which is set accurately usinga bandgap regulator to generate a reference current and using currentmirrors to set the pump current. The step sizes used in the illustratedembodiment are shown in FIG. 72. Of course, other step sizes can beemployed, as desired, and various numbers of different sized steps canbe employed.

The time used for the calculations for the coarse and medium cases is 40ns, a typical value for the starting period of the oscillator. 26.2 nsis used for the medium fine and fine cases because these steps occurwhen the oscillator is close to its final frequency.

The course and medium steps are controlled by the Puf1 and Puf2 outputsof the start-up circuit. More particularly, in the illustratedembodiment, the course steps are controlled by the PumpUpFast output ofthe start-up circuit “dcr_startup” shown in the detailed schematicdrawings, and the medium step is controlled by the PumpUpMed output ofthe start-up circuit “dcr_startup” shown in the detailed schematicdrawings. The medium fine step is also controlled by the PumpUpMedsignal but the step size is reduced when the SDD (start data decode)signal is asserted indicating the oscillator is within a few percent ofits final value. The fine step is controlled by the state machine and isused to “close in” on the final value.

While this charge pump and loop filter configuration is advantageous forimplementation on an integrated circuit, other configuration arepossible. For example, simple RC filters can be employed.

Details of Transmit Frequency Derivation from Incoming Data

The illustrated embodiment has a loop filter including capacitors onrespective control nodes “OUTN” and “OUTP” (shown in FIGS. 8.0504AA-EE)of the voltage controlled oscillator “vco.” In the illustratedembodiment, the loop filter capacitor on the control node “OUTP” isdefined by a plurality of capacitors in parallel, and the loop filtercapacitor on the control node “OUTN” is defined by a plurality ofcapacitors in parallel. The voltage on the respective control nodes isindicative of the frequency at which the voltage controlled oscillator“vco” is oscillating. After an entire incoming message has been receivedby the receiver “rx,” the control nodes and the capacitors on thecontrol nodes are isolated from driving circuitry. The control voltageis thus stored in analog form on the capacitors, and the voltagecontrolled oscillator “vco” continues to oscillate at the recoveredfrequency. The length of time that the voltage stored on the capacitorsis valid depends on leakage currents that can charge or discharge thecapacitors over time.

In the illustrated embodiment, such leakage currents are minimized byminimizing n+ and p+ active areas on the control node, and by minimizingdrain to source voltages on devices connected to the control nodes. Thevalues for the respective capacitors are chosen, in conjunction withloop filter requirements, to hold the control voltages for as long aspossible as required before the device 12 transmits a reply to thereceived radio frequency command. This amount of time is approximatelyseveral hundred milliseconds in the illustrated embodiment.

The output frequency of the voltage controlled oscillator can bemultiplied up to generate a carrier frequency for the transmitter, asdescribed elsewhere, or can be divided down to generate tones for FSK(frequency shift keyed) transmission or DPSK (differential phase shiftkeyed) transmission depending on what form of transmission is selectedfor the transmitter “tx.”

In one embodiment, only one control node is employed; however, in theillustrated embodiment, a differential control node scheme is employedinvolving two control nodes “OUTN” and “OUTP.” Therefore, in theillustrated embodiment, a capacitor is provided on each control node,and control voltages are stored in analog form on these two capacitors.

FIGS. 8.0504AA-EE provide a circuit drawing of the control voltagegenerator “dcr_vcocontrol.” The control voltage generator shows thecontrol nodes for the voltage controlled oscillator. The control voltagegenerator is a differential circuit. The control nodes are shown on theright edge of FIGS. 8.0504AA-EE as “OUTP” and “OUTN,” where “OUTN” isactually tied to the band gap voltage, which is approximately 1.2 Volts.“OUTP” is the node that is pumped up to adjust frequency. The controlvoltage generator includes step size generators shown on the left halfof FIGS. 8.0504AA-EE. The steps are achieved by conducting a current tothe capacitor on the control node for a prescribed length of time. For alarge step, a large current is applied to this capacitor. For a smallstep, a smaller current is applied to this capacitor. The capacitor onthe control node “OUTP” is defined by ten capacitors in parallel in theillustrated embodiment.

A similar capacitor, defined by ten capacitors in parallel, is providedon the other control node “OUTN.”

Four different size currents are generated by fine, medium fine, medium,and coarse step generators “dcr_finestepgen,” “dcr_medfinestepgen,”“dcr_medstepgen,” and “dcr_coarsestepgen” respectively. The currents areeither steered to the control capacitor on the control node or away fromthe capacitor, depending on whether there is a pump up or pump downcommand.

FIGS. 8.050401AA-CK provide a circuit drawing showing constructiondetails of the coarse step generator “dcr_coarsestepgen.” The coarsestep generator includes a plurality of current mirrors.

FIGS. 8.050402AA-CJ provide a circuit drawing showing constructiondetails of the medium step generator “dcr_medstepgen.” The medium stepgenerator includes a plurality of current mirrors.

FIGS. 8.050403AA-BI provide a circuit drawing showing constructiondetails of the medium fine step generator “dcr_medfinestepgen.” Themedium fine step generator includes a plurality of current mirrors.

FIGS. 8.050404AA-BB provide a circuit drawing showing constructiondetails of a fine step controller “dcrfinestepctrl.”

FIGS. 8.050405AA-EJ provide a circuit drawing showing constructiondetails of the fine step generator “dcr_finestepgen.”

FIGS. 8.0505AA-EF provide a circuit drawing of the voltage controlledoscillator “dcr_vco.” The voltage controlled oscillator “dcrvco” is afour stage ring oscillator with differential stages. The voltagecontrolled oscillator includes an OTA (operational transconductanceamplifier) shown on the left side of FIGS. 8.0505AA-DE. The OTA gives alinear relationship between the voltage differential at its inputs andthe current at its output. The voltage controlled oscillator furtherincludes current mirrors which mirror the current at the output of theOTA to drive the voltage controlled oscillator to change its frequency.The previously discussed control nodes (“OUTN” and “OUTP” of FIGS.8.0504AA-EE) are shown coming in on the left side FIGS. 8.0505AA-DE,labelled as “INN” and “INP.” The voltage controlled oscillator furtherincludes, at its output, a comparator type circuit that provides digitallevels for the output of the voltage controlled oscillator “dcr_vco.”

FIG. 8.0506AA-AB provide a circuit drawing of a clock generator“dcr_rxclkgen.” Different frequencies are needed for differentfunctions. The clock generator provides outputs at differentfrequencies. For example, the clock generator provides an output“PROCCLK” (for the processor), an output “CHIPCLK” (chip clock); andoutputs “PLLCLKP” and “PLLCLKN” for the clock that drives the statemachine. The clock generator “dcr_rxclkgen” has an input “LOWRATE” forlow rate which is a signal indicative that the chip is in low rate andcan expect data to come in at a chip rate of one-half the normal chiprate. The loop is adjusted in a manner such that the frequency of“FMASTER” does not change regardless of whether the chip is in high rateor low rate. However, the clock “CHIPCLK” for the integrated circuit 16is half as fast in low rate, and it takes twice as long to get data inas it would to get the same amount of data in at the high rate.

FIG. 8.050601 provides a circuit drawing showing construction details ofa flip-flop “dcr_rxclkgenff” included in the clock generator.

FIGS. 8.0507AA-AB provide a circuit drawing of a non-overlapping clockgenerator “dcr_clkgen.” The non-overlapping clock generator receives asinputs true and compliment clock signals “ClkInP” and “ClkInN” andprovides buffered true and compliment clock signals “ClkOut” and“ClkOutN.” The non-overlapping clock generator buffers the true andcompliment clock signals “ClkInP” and “ClkInN” in such a way that before“ClkOut” can go high, “ClkOutN” must be low, and then at the end of thatcycle, before “ClkOutN” can go high, “ClkOut” must be low. Any overlapbetween the two clocks occurs when they are both low. They are neverboth high at the same time. This is quite commonly required in manycircuits throughout the integrated circuit 16 where shift register typetechniques are used, and one stage passes information to another.Non-overlapping clocks are required for such functions.

The circuit of FIGS. 6AA-EK further includes a transmitter “tx.” Thetransmitter “tx” is capable of transmitting using different modulationschemes, and the modulation scheme is selectable by the interrogator.More particularly, if it is desired to change the modulation scheme, theinterrogator sends an appropriate command via radio frequency. Thetransmitter can switch between multiple available modulation schemessuch as Frequency Shift Keying (FSK), Binary Phase Shift Keying (BPSK),Direct Sequence Spread Spectrum, On-Off Keying (OOK), AmplitudeModulation (AM), and Modulated Backscatter (MBS).

The output responses are included in packets that contain, in order oftransmission, a preamble, a Barker code, and the reply data.

In one embodiment, each bit of the radio frequency reply sent by thedevice 12 is modulated using a pseudo noise (PN) sequence for directsequence spread spectrum communication. The sequence is generated inpart by a linear feedback shift register “pngshr” having a plurality ofregisters “pngsreg.” In one embodiment, the linear feedback shiftregister is in the form [5,2] which means that the input to the firstregister is the result of combining the output of the fifth register bythe exclusive-OR with the output of the second register. This producesthirty-one states. In one embodiment, the linear feedback shift registeris in the form [6,1] for a sixty-three chip sequence. In anotherembodiment, the linear feedback shift register is in the form [8,4,3,2]for a two hundred and fifty-five chip sequence. In a preferredembodiment, the shift register is selectable between multiple of theabove forms. In the form [6,1], the input to the first of six registersis the result of combining the output of the sixth register byexclusive-OR with the output of the first register. In the form[8,4,3,2], the input to the first of eight registers is the result ofcombining the outputs of registers eight, four, three, and two byexclusive-OR. The sixty-three chip output sequence requires less timefor signal synchronization than the two hundred and fifty-five chipsequence. However, the two hundred and fifty-five chip output sequenceprovides better performance in systems having poor signal to noiseratio.

FIGS. 8.06AA-ED provide a circuit drawing of the transmitter “tx.” FIGS.8.06AA-ED show a transmitter PLL “txpllfsyn,” a test mode data selector“txdatasel,” a BPSK modulation driver “txbpsk,” a frequency doubler“txdoubler,” a second frequency doubler “txdoubler2,” a transmitterpower amp “txpoweramp,” a transmitter bias generator “txbias,” and amodulated backscatter transmitter “txmbs.” FIGS. 8.06AA-ED actuallyshows two different transmitters. Much of FIGS. 8.06AA-ED illustratescircuitry employed for an active transmitter which is used in accordancewith an alternative embodiment of the invention, but not in accordancewith the preferred embodiment. FIGS. 8.06AA-ED also illustrate themodulated backscatter transmitter “txmbs” that is employed in apreferred embodiment. The active transmitter will be discussed first.

In embodiment where the active transmitter is employed, the activetransmitter operates by taking the “FMASTER” clock that was recoveredfrom the incoming data stream and using a phase locked loop “txpllfsyn”(an analog phase locked loop in the illustrated embodiment) to multiplyup the frequency. In the illustrated embodiment, the frequency ismultiplied up by a factor of sixteen from 38 MHZ to 610 MHZ. The phaselocked loop includes an oscillator that generates eight phases which are45° out of phase with respect to each other. The eight phases generatedby the oscillator: are applied to first doubler circuits “txdoubler” and“txdoubler2” in order to generate the proper phased outputs at doublethe frequency that then again serve as inputs to the other doublercircuit. The active transmitter further includes a transmitter power amp“txpoweramp.” The transmitter power amp includes the other doubler thatreceives the outputs of the first doubler circuits “txdoubler” and“txdoubler2.” Capability for several different modulation techniques isprovided for the active transmitter. One such modulation technique isBPSK where the phase of the carrier (2.44 GHz in the illustratedembodiment) is inverted to indicate a bit change. Another suchmodulation technique is amplitude modulation (AM). In the illustratedembodiment, 100% modulation, or on/off keying, is employed with theamplitude modulation.

FIGS. 8.0601AA-BB provide a circuit drawing of the transmitter phaselocked loop “txpllfsyn.” The phase locked loop “txpllfsyn” includes avoltage controlled oscillator “txvco” that receives an analog tunevoltage and provides an output frequency in accordance with the analogtune voltage. The phase locked loop further includes a divider“txdivider” which receives the output signal of the voltage controlledoscillator “txvco” and divides the frequency of the output of theoscillator “txvco” by a factor of sixteen. It will be understood thatthis division ratio of sixteen is for an exemplary embodiment, and thescope of the present invention encompasses other division ratios. Thephase locked loop “txpllfsyn” includes a phase/frequency detector“txpfdet.” At the phase/frequency detector “txpfdet,” the output of thedivider “txdivider” is compared to the signal received at the referenceinput of the detector, which reference input, in accordance with oneembodiment, is the signal “FMASTER” recovered from the incoming datastream. The phase/frequency detector “txpfdet” compares the fed backsignal (i.e., having a frequency of the voltage controlled oscillator“txvco” divided by sixteen) with the signal received at the referenceinput and puts out a pump up signal “PU” or pump down signal “PD” inaccordance with phase and frequency difference therebetween. The phaselocked loop “txpllfsyn” further includes a charge pump “txchgpump.” Thepump up signal “PU” or pump down signal “PD” put out by thephase/frequency detector “txpfdet” drive the charge pump “txchgpump.”The phase locked loop “txpllfsyn” further includes a loop filter“txloopfilter” that receives an output signal from the charge pump“txchgpump” and filters this output signal for providing the tunevoltage for controlling the voltage controlled oscillator “txvco.” Thefilter “txloopfilter” removes transients and establishes loop dynamics,i.e. responsiveness, of the resulting phase locked loop.

Again, the voltage controlled oscillator provides an output signalhaving a frequency proportional to the tune voltage received at itsinput. When the phase locked loop is locked, the frequency and phase ofthe signal fed back to the phase/frequency detector is equal to thefrequency and phase of the reference input signal. Therefore, the outputfrequency of the voltage controlled oscillator “txvco” is equal to Ntimes the frequency of the reference signal, where N is equal to thedivision factor of the divider. For the exemplary embodiment describedabove, N is equal to sixteen and the output frequency of the voltagecontrolled oscillator “txvco” is equal to sixteen times the frequency ofthe reference signal, e.g. 16×38.15 MHZ=610.45 MHZ. By providing variousoutput taps distributed along a ring topology of the voltage controlledoscillator “txvco,” output signals of different phase relationships (butof equal frequency) are obtained from the voltage controlled oscillator“txvco.” In a preferred embodiment, eight separate output taps from thevoltage controlled oscillator “txvco” provide eight different outputsignals having substantially 45° differences in phase therebetween,e.g., 0°, 45°, 90°, 135°, 180°, 225°, 270°, and 315°. Thus, in theillustrated embodiment, the voltage controlled oscillator “txvco”generates eight phases spaced 45°.

FIGS. 8.060101AA-BB provide a circuit drawing showing constructiondetails of the phase/frequency detector “txpfdet.” The phase/frequencydetector puts out a pump up signal “PU” or pump down signal “PD”, anddrives the charge pump.

FIGS. 8.060102AA-BB provide a circuit drawing showing constructiondetails of the charge pump “txchgpump.” The charge pump drives the loopfilter.

FIGS. 8.060103AA-CB provide a circuit drawing showing constructiondetails of the loop filter “txloopfilter.”

FIGS. 8.060104AA-DC provide a circuit drawing showing constructiondetails of the transmitter voltage controlled oscillator “txvco.” Thevoltage controlled oscillator “txvco” generates eight phases spaced 45°.

Details of CMOS High Freguencv VCO Stage

The voltage controlled oscillator “txvco” comprises a ring oscillatorhaving four stages. FIG. 32 is a simplified schematic illustrating onestage 104. Four such stages are connected in a chain, with the outputsof the chain connected to the inputs of the chain, to define the ringoscillator. The stage 104 includes a p-channel transistor 105 having agate defining a control node “V control,” having a source connected to asupply voltage “V+,” and having a drain; and a p-channel transistor 106having a gate connected to the control node “V control,” having a sourceconnected to the supply voltage “V+,” and having a drain. The stage 104further includes an n-channel transistor 107 having a gate defining aninput “IN P,” a drain connected to the drain of the transistor 105 anddefining a node “B,” and a source; and an n-channel transistor 108having a gate defining an input “IN N,” a drain connected to the drainof the transistor 106 and defining a node “A,” and a source. The stage104 further includes an ideal current source 109 connected to thesources of the transistors 107 and 108 and directing current from thesources of the transistors 107 and 108 to ground. The stage 104 furtherincludes a resistor 110 connected between the voltage “V+” and the drainof the transistor 107, and a resistor 111 connected between the voltage“V+” and drain of the transistor 108. The stage 104 further includes asource follower 112 including an n-channel transistor 113 having a gateconnected to the node “A,” having a drain connected to a supply voltage“V+,” and having a source defining an output “OUT P”; and an idealcurrent source 114 connected to the source of the transistor 113 anddirecting current from the source of the transistor 113 to ground. Thestage 104 further includes a source follower 115 including an n-channeltransistor 116 having a gate connected to the node “B,” having a drainconnected to the supply voltage “V+,” and having a source defining anoutput “OUT N”; and an ideal current source 117 connected to the sourceof the transistor 116. A source follower is a circuit where the signalat the source terminal of a transistor is approximately equal to thesignal at the gate of the transistor. The source followers 112 and 115are provided in the stage 104 to provide the necessary drive for theoutputs “OUT P” and “OUT N” to drive a load. More particularly, theoutputs “OUT P” and “OUT N” drive amplifiers that drive frequencydoublers described elsewhere herein. Nodes “A” and “B” are connected toanother stage in the chain (e.g., by connecting the nodes “A” and “B” toinputs “IN P” and “IN N” of a subsequent stage).

The ideal current source 109 drives a current “IBIAS,” and the values ofthe resistors 110 and 111 and of “IBIAS” are chosen such thattransistors 107 and 108 are in saturation. More particularly, the valuesof the resistor 110 and the current “IBIAS” are chosen such that thevalue of the resistance of resistor 110 multiplied by the current“IBIAS” is less than a maximum voltage (e.g. 800 mV) to cause saturationof transistor 107. In the illustrated embodiment, resistors 110 and 111have the same resistance value. The resistors 110 and 111 are made fromn-well, n+, p+, or polysilicon depending on the process used tomanufacture the integrated circuit 16. Parasitic capacitance on nodes Aand B is minimized by compact arrangement of the components of the stage104. Computer software, such as from Cadence, can also be employed toreduce parasitic capacitance.

The stage 104 provides a differential amplifier capable of switching ata very high frequency. The switching frequency is adjustable byadjusting the voltage at control node “V control.” More particularly, asthe voltage at the control node “V control” decreases, the p-channeltransistors 105 and 106 become more conductive, and there is lessimpedance between the supply voltage “V+,” and the drains of thetransistors 107 and 108. This provides for a faster switching rate.There is a linear change in frequency versus the voltage at the controlnode “V control” for at least some range of voltages.

FIGS. 8.06010401AA-DC provide a circuit drawing showing constructiondetails of a stage “txvcostage” included in the transmitter voltagecontrolled oscillator. FIG. 8.0601040101 is a layout plot showing howthe components of the stage 104 of FIG. 32 are laid out. Moreparticularly, FIG. 8.0601040101 actually shows four stages. FIG.8.0601040101 shows locations 400 defining resistors; a line 402providing VREG (V+ of FIG. 32); locations 404 defining source followersof FIG. 32; a location 406 defining input “IN P” of FIG. 32; a location408 defining input “IN N” of FIG. 32; locations 410 defining then-channel differential pair and the current source of FIG. 32; locations412 defining the p-channel devices of FIG. 32; a line 414 providing thecontrol voltage “V control” of FIG. 32; and 915 MHZ option capacitors416.

FIGS. 8.060105AA-DD provide a circuit drawing showing constructiondetails of the divider “txdivider.”

FIGS. 8.06010501AA-AB provide a circuit drawing showing constructiondetails of a flip-flop “txdivtff” included in the divider.

FIGS. 8.0602AA-AB provide a circuit drawing showing construction detailsof a test mode data selector “txdatasel.”

FIGS. 8.0603AA-AB provide a circuit drawing showing construction detailsof a BPSK modulation driver “txbpsk.”

Details of Freguency Doubler

Analog multipliers are known in the art. An analog multiplier includestwo inputs, and includes an output providing a signal that isrepresentative of a multiplication of one of the inputs with the otherof the inputs. One known analog multiplier is known as a Gilbertmultiplier cell. For a detailed discussion of Gilbert cells, see FourQuadrant Multiplier, B. Gilbert, IEEE Journal of Solid State Circuits,1968, pp. 365-373. Such Gilbert multiplier cells are also described indetail in Analysis and Design of Analog Integrated Circuits, Paul R.Gray and Robert G. Meyer, Third Edition, 1993, John Wiley & Sons, Inc.,pp. 667-681. Such Gilbert multiplier cells include two cross-coupled,emitter-coupled pairs of bipolar junction transistors in series with anemitter coupled pair of bipolar junction transistors. A Gilbertmultiplier cell employing bipolar junction transistors produces anoutput that is the hyperbolic tangent of two input voltages. This isbecause a characteristic of bipolar junction transistors is exponentialnon-linearity. If the input voltages are sufficiently low, thehyperbolic tangent function can be approximated as linear, and thecircuit behaves as a multiplier which multiplies together the two inputvoltages.

The multiplier cell originally developed by Gilbert employed bipolarjunction transistors. It is also known to employ MOS transistors toproduce a Gilbert multiplier cell. See, for example, Analog IntegratedCircuits for Communication, Principles, Simulation and Design, Donald O.Pederson and Kartikeya Mayaram, Kuwer Academic Publishers, ThirdPrinting, 1994, pp. 431-433.

FIG. 34 illustrates a frequency doubler circuit 119 that includes aGilbert cell 120. The Gilbert cell 120 includes a pair 122 defined bytransistors 124 and 126. The Gilbert cell 120 further includes a pair128 defined by transistors 130 and 132. The transistors 124 and 126 havesources that are connected together. Thus, the pair 122 is a sourcecoupled pair. The transistors 130 and 132 have sources that areconnected together. Thus, the pair 128 is a source coupled pair.

The transistors 126 and 130 have gates that are connected together todefine a first input node. The transistors 124 and 132 havegates thatare connected together to define a second input node. The transistors124 and 130 have drains that are connected together, and the transistors126 and 132 have drains that are connected together (shown as acrisscross pattern in FIG. 34).

The Gilbert cell 120 further includes another pair 134 includingtransistors 136 and 138 having sources coupled together. Thus, the pair134 is a source coupled pair. The pair 134 is in series with the pairs122 and 128. More particularly, the transistor 136 has a drain connectedto the sources of the transistors 124 and 126, and the transistor 138has a drain connected to the sources of the transistors 130 and 132. Thetransistor 138 has a gate defining a third input node, and thetransistor 136 has a gate defining a fourth input node.

The Gilbert cell 120 further includes an ideal current source 140driving current from the sources of the transistors 136 and 138 toground. The frequency doubler 119 further includes a resistor 142connected between the drain of the transistor 124 and a voltage, and aresistor 144 connected between the drain of the transistor 132 and thevoltage. The resistors 142 and 144 define loads for current steeringthat produces output voltage swings.

For low amplitude signals, the Gilbert cell 120 provides an outputbetween the drain of the transistor 124 and the drain of the transistor132 that is an analog multiplication of a first input signal appliedbetween the first and second input nodes, by a second input signalapplied between the third and fourth input nodes.

It is known to use a Gilbert cell to multiply together sine waves ofdifferent phases to produce a doubled frequency (FIGS. 34 and 35). Thisis based on a known trigonometric relationship:

sin 2θ=2 sin θ cos θ

Signals that are 180° apart are applied to the first and second inputnodes, and a phase shifter produces 90° shifted signals that are appliedto the third and fourth input nodes. However, in such embodiments, anintegrator is required, and the phase shifter is required to be feedbackcontrolled, because slight errors in the required 90° phase shift wouldotherwise cause the output signals to have different average values anddifferent amplitudes as shown in FIG. 33. FIG. 33 is a waveform diagramillustrating the effect of errors in frequency doubler circuits thatnecessitates correction, such as by using an integrator and feedback.FIG. 34 is a circuit schematic illustrating a frequency doubler circuitthat employs an integrator and feedback to solve the problem illustratedin FIG. 33. FIG. 35 is a waveform diagram illustrating input and outputwaves created and employed by a frequency doubler circuit such as theone shown in FIG. 34.

It is desirable to avoid the need for feedback. Frequency multipliercircuits employing feedback are susceptible to being disturbed. Forexample, if substrate noise or an adjacent line switches and causes ashift at the integrator, the output will be distorted from the desiredoutput until the integrator has a chance to recover. The integrator cantake a long time to recover. Therefore, it is desirable to eliminatefeedback loops from a frequency multiplier.

FIG. 36 is a circuit schematic illustrating a symmetric frequencydoubler circuit 146 that does not require an integrator and feedback tosolve the problem illustrated in FIG. 33. The frequency doubler circuitof FIG. 36 creates and employs waveforms such as those shown in FIG. 35.

The frequency doubler circuit 146 includes a first Gilbert cell 148, anda second Gilbert cell 150 coupled to the first Gilbert cell 148.

The first Gilbert cell 148 includes a pair 152 defined by transistors154 and 156. The transistors 154 and 156 have sources that are connectedtogether. Thus, the pair 152 is a source coupled pair. The Gilbert cell148 further includes a pair 158 defined by transistors 160 and 162. Thetransistors 160 and 162 have sources that are connected together. Thus,the pair 158 is a source coupled pair.

The transistors 156 and 160 have gates that are connected together todefine a first input node 163. The transistors 154 and 162 have gatesthat are connected together to define a second input node 165. Thetransistors 154 and 160 have drains that are connected together, and thetransistors 156 and 162 have drains that are connected together (shownas a crisscross pattern in FIG. 36).

The Gilbert cell 148 further includes another pair 164 includingtransistors 166 and 168 having sources coupled together. Thus, the pair164 is a source coupled pair. The pair 164 is in series with the pairs152 and 158. More particularly, the transistor 166 has a drain connectedto the sources of the transistors 154 and 156, and the transistor 168has a drain connected to the sources of the transistors 160 and 162. Thetransistor 168 has a gate defining a third input node 169, and thetransistor 166 has a gate defining a fourth input node 171.

The Gilbert cell 148 further includes an ideal current source 170driving current from the sources of the transistors 166 and 168 toground. The frequency doubler 146 further includes a resistor 172connected between the drain of the transistor 154 and a voltage, and aresistor 174 connected between the drain of the transistor 162 and thevoltage. The resistors 172 and 174 define loads for current steeringthat produces output voltage swings.

The second Gilbert cell 150 includes a pair 182 defined by transistors184 and 186. The transistors 184 and 186 have sources that are connectedtogether. Thus, the pair 182 is a source coupled pair. The Gilbert cell150 further includes a pair 188 defined by transistors 190 and 192. Thetransistors 190 and 192 have sources that are connected together. Thus,the pair 188 is a source coupled pair.

The transistors 186 and 190 have gates that are connected together todefine a first input node 193 of the second Gilbert cell 150. Thetransistors 184 and 192 have gates that are connected together to definea second input node 195 of the second Gilbert cell 150. The transistors184 and 190 have drains that are connected together, and the transistors186 and 192 have drains that are connected together (shown as acrisscross pattern in FIG. 36).

The Gilbert cell 150 further includes another pair 194 includingtransistors 196 and 198 having sources coupled together. Thus, the pair194 is a source coupled pair. The pair 194 is in series with the pairs182 and 188. More particularly, the transistor 196 has a drain connectedto the sources of the transistors 184 and 186, and the transistor 198has a drain connected to the sources of the transistors 190 and 192. Thetransistor 198 has a gate defining a third is input node 199, and thetransistor 196 has a gate defining a fourth input node 201.

The Gilbert cell 150 further includes an ideal current source 200driving current from the sources of the transistors 196 and 198 toground.

The outputs of the second Gilbert cell are connected to the outputs ofthe first Gilbert cell. More particularly, the drain of the transistor184 is connected to the drain of the transistor 154 and the drain of thetransistor 192 is connected to the drain of the transistor 162.

The first input node 193 of the second Gilbert cell 150 is connected tothe fourth input node 171 of the first Gilbert cell 148. The third inputnode 199 of the second Gilbert cell 150 is connected to the second inputnode 165 of the first Gilbert cell 148. The fourth input node 201 of thesecond Gilbert cell 150 is connected to the first input node 163 of thefirst Gilbert cell 148.

In operation, a first sinusoidal signal is applied to the second inputnode 165 of the first Gilbert cell 148. A second sinusoidal signal, 180°out of phase with the first sinusoidal signal, is applied to the firstinput node 163 of the first Gilbert cell 148 (and to the fourth inputnode of the second Gilbert cell 150). A third sinusoidal signal, 90° outof phase with the first sinusoidal signal, is applied to the secondinput node 195 of the second Gilbert cell 150. A fourth sinusoidalsignal, 270° out of phase with the first sinusoidal signal, is appliedto the first input node of the second Gilbert cell 150. Thisrelationship of phases on the inputs to the first and second Gilbertcells causes the output to be symmetrical so that the problem of FIG. 33is avoided without the need for feedback. Even with slight errors inphases between the input signals, a symmetrical output is produced.

Generally speaking, each Gilbert cell adds current from bottomtransistors to top transistors through the resistor loads to form outputvoltages. In the illustrated embodiment, a phase arrangement applied tothe upper Gilbert cell is generally reversed for the bottom Gilbert cellso undesirable offsets cancel each other.

In one embodiment, the second, third, and fourth input sinusoidalsignals are derived from the first input sinusoidal signal using asimple four stage differential oscillator.

A doubled frequency is thus obtained at the outputs, which are definedat the drain of the transistor 154 and the drain of the transistor 162,without the need for an integrator and feedback.

FIGS. 8.0604AA-AB provide a circuit drawing of the frequency doubler“txdoubler.” The frequency doubler circuit “txdoubler” includes adoubler core “txfdbl” having two tiers of transistors. The two tiers oftransistors are shown in FIGS. 8.060401AA-FE as being a top tier and abottom tier. The frequency doubler requires different levels dependingon whether the top tier of transistors or bottom tier of transistors aredriven by a particular phase. The frequency doubler “txdoubler”therefore includes driver amplifiers “txfdbldrv” which provide thatlevel shifting. There is no intended phase shift introduced by thedriver amplifiers.

FIGS. 8.060401AA-FE provide a circuit drawing of the frequency doublercore “txfdbl.” The frequency doubler core “txfdbl” includes levelshifting circuitry. The level shift is a little level shift, and isaccomplished by a resistor and capacitor shown at the top of the rightstack (FIG. 8.060401AD). The level shift is performed in order to adjustoutput levels down in voltage because this frequency doubler core drivesanother frequency doubler.

FIGS. 8.0605AA-AB provide a circuit drawing of the frequency doubler“txdoubler2.” The frequency doubler “txdoubler2” is substantiallysimilar to the first frequency doubler “txdoubler.” The main differencehas to do with the bias arrangements for the driver amps and for thedoubler core. In an alternative embodiment, the first and secondfrequency doublers “txdoubler” and “txdoubler2” are identical.

FIGS. 8.060501AA-CD provide a circuit drawing showing constructiondetails of the doubler driver amplifier “txfdbldrv.”

FIGS. 8.060502AA-CD provide a circuit drawing showing constructiondetails of a second doubler driver amplifier “txfdbldrv2” included inthe frequency doubler “txdoubler2.” The second doubler driver amplifier“txfdbldrv2” include a bias diode. The doubler driver amplifier“txfdbldrv2” includes circuitry (the criss-cross configuration in FIGS.8.060502AA-CD) where bi-phase (binary phase shift keying) modulation isperformed. This is where a phase can be switched from one side toanother by the state of two inputs “BPMODINP” and “PMODINN.” Thus, aphase reversal can be accomplished in this circuit.

FIGS. 8.060503AA-FE provide a circuit drawing of a frequency doublercore “txfdbl2.” The frequency doubler core “txfdbl2” is substantiallyidentical to the frequency doubler core “txfdbl” except for the biasingtransistors.

Details of Single Antenna Receiver and Active Transmitter

FIG. 50 provides a simplified circuit schematic showing the antenna 44being shared by the active transmitter and the Schottky diode detector84. The Schottky diode detector 84 was described above in detail inconnection with FIG. 29, like reference numerals indicating likecomponents.

The detector 84 includes a Schottky diode 86 having an anode connectedto the antenna 44 and having a cathode. The detector 84 further includesan ideal current source 88 connected to the cathode of the Schottkydiode 86 and driving current through the antenna and Schottky diode 86in the direction from the anode to the cathode. The detector 84 furtherincludes a capacitor 90 connected between the cathode of the Schottkydiode 86 and ground and providing a radio frequency short to ground. Thedetector 84 further includes a capacitor 92 having a first terminalconnected to the cathode, having a second terminal defining an output ofthe detector 84, providing an AC short to video frequency, and definingthe output of the detector 84.

The active transmitter is described elsewhere herein, and is illustratedas a block 330 in FIG. 50.

The antenna is a loop antenna and has one end connected to a biasvoltage (Vdd) and has another end connected to the anode of the Schottkydiode 86.

The transmitter has an antenna output (or RF output) 332, and thedetector 84 has an antenna input (or RF input) 334. In the illustratedembodiment, the integrated circuit 16 having the transmitter 330 anddetector 84 includes a contact connected to the antenna output 332 andaccessible from outside the IC package; and a contact connected to theantenna input 334 and accessible from outside the IC package. Thesecontacts are connected together by a short outside the package. Thisprovides for flexibility in that different antenna configurations arepossible, separate antennas can be used for the detector 84 andtransmitter 330, if desired, an external amplifier can be used toamplify the output of the transmitter 330, etc.

The detector and transmitter do not operate simultaneously.

In one embodiment, the integrated circuit 16 further includes a pull uptransistor 336 connected to the cathode of the Schottky diode 86 andconfigured to connect the cathode to the bias voltage (Vdd) when thetransmitter is operating. The pull up transistor 336 can be included ifnecessary so the detector does not interfere with the transmitter 330while the transmitter 330 is transmitting.

By using a common antenna for the active transmitter and the Schottkydiode detector, space savings are achieved.

The active transmitter 330 is shown in greater detail in FIG. 51. Theactive transmitter includes a differential pair 338 of transistorsdriven by the frequency doubler. The function of the differential pair338 is to steer current to the antenna 44 or away from the antenna 44.If bi-phase modulation is employed, the differential pair 338 steers onephase or the other phase to the antenna 44. More particularly, ifbi-phase modulation is employed, then a signal on line “ENABLEAM” (seeFIGS. 8.06AA-ED, and 8.0605AA-AB to 8.0608AA-BB) is low and the leftmostof three current steering transistors (sources connected to the currentsource) is off because its gate is low. Current is then steered to theantenna by the transistor shown on the right. Its phase is determined inan earlier stage. The other phase is present in the middle transistor.When data is reversed the current phases switch sides in response to theearlier stage. If amplitude modulation is employed, the differentialpair either sends current to the antenna 44, or is sends none to provideon/off keying. More particularly, in the amplitude modulation mode, asignal on line “ENABLEAM” is high and current is steered to the antennaby the transistor shown to the right if a signal on line “AMDATA” ishigh, and the current is steered to Vdd (not to the antenna) if thesignal on line “AMDATA” is low.

FIGS. 8.0606AA-IE provide a circuit drawing of a transmitter poweramplifier “txpoweramp.” The transmitter power amplifier includes afrequency doubler, shown in the left half of FIGS. 8.0606AA-IE. In theillustrated embodiment, the frequency doubler receives inputs at 1.22GHz, and provides outputs at 2.44 GHz. The transmitter power amplifierincludes the differential pair of transistors, shown on the right sideof FIGS. 8.0606AA-IE, driven by the frequency doubler. The differentialpair steers current to the antenna or away from the antenna, asdescribed above. If bi-phase modulation is employed, the differentialpair steers one phase or the other phase to the antenna. If amplitudemodulation is employed, the differential pair either sends current tothe antenna, or it sends none to provide on/off keying.

FIGS. 8.0607AA-JJ provide a circuit drawing of a transmitter biasgenerator “txbias.” The transmitter bias generator includes variouscurrent mirrors in order to provide the proper bias currents to thevarious blocks of the transmitter “tx.”

Details of Single Antenna Receiver and Backscatter Transmitter

FIG. 52 provides a simplified circuit schematic showing an antenna 350being shared by the backscatter transmitter and the Schottky diodedetector 84, in a manner similar to the antenna sharing possibilitydescribed in connection with FIGS. 50-51. The Schottky diode detector 84was described above in detail in connection with FIG. 29, like referencenumerals indicating like components.

In the illustrated embodiment, the antenna 350 is a loop antenna and hasone end connected to a bias voltage (Vdd) and has another end connectedto the detector 84 via a detector input illustrated as RXANT in FIG. 52.For antenna sharing with a backscatter transmitter, capacitors 352 and354 external of the integrated circuit 16 are employed, as illustratedin FIG. 52, to isolate the antenna from the backscatter antenna driverwhen the detector is using the antenna.

The detector and transmitter do not operate simultaneously.

By using a common antenna for the backscatter transmitter and theSchottky diode detector, space savings are achieved.

In an alternative embodiment shown in FIG. 53, a single antenna 350 isshared by the detector 94 (shown in FIG. 30 and described elsewhereherein) and a backscatter transmitter. An n-channel transistor 356 isprovided having power electrodes connected to opposite ends of theantenna, and having a control electrode connecting to transmittermodulation circuitry. The control electrode is held low when the antennais being used by the receiver.

FIGS. 8.0608AA-BB provide a circuit drawing of a modulated backscattertransmitter “txmbs.” The modulated backscatter transmitter “txmbs”includes circuitry that creates non-overlapping drive signals. Themodulated backscatter transmitter “txmbs” includes primary antenna ports“BS1” and “BS2.” Each of these antenna ports is intended to be connectedto one-half of a dipole antenna having a length appropriate for thetransmission frequency. In the illustrated embodiment, the halves of thedipole antenna have respective sizes appropriate for 2.44 Ghz. Thehalves of the dipole antenna are not included on the integrated circuit16, in the illustrated embodiment, but are instead provided “off chip.”Other antenna arrangements are possible.

The modulated backscatter transmitter “txmbs” further includes ann-channel transistor marked 900 micron in FIGS. 8.0608AA-BB, and twon-channel pull-up transistors marked 100 micron and respectivelyconnected between a voltage vdd! and the 900 micron transistor. When thegate of the transistor marked 900 micron is high, then the two dipolehalves are shorted together with a fairly low impedance (e.g., on theorder of 15 Ohms, plus any bond wire impedance that might be presentdepending on how the device is packaged). The antenna becomessubstantially similar to a single half-wavelength antenna. In abackscatter mode, when the two halves of the antenna are shortedtogether, the antenna reflects a portion of the power being transmittedby the interrogator. In the other state, the gate of the 900 microntransistor is low. The 900 micron transistor is then off, but the two100 micron transistors that pull up the voltage vdd! are on, liftingantenna ports “BS1” and “BS2” both up to a voltage of vdd! minus ann-channel Vt. The two antenna ports “BS1” and “BS2” are then isolatedfrom each other by an open circuit. This isolation changes the radarcross-section of the dipole antenna dramatically from when the twohalves are shorted together. The antenna becomes substantially similarto two quarter wavelength antennas. In a Backscatter mode, when the twohalves of the antenna are isolated, the antenna reflects very little ofthe power transmitted by the interrogator.

The modulated backscatter transmitter “txmbs” further includescross-coupled circuitry shown near the middle in FIGS. 8.0608AA-BB. Thecross-coupled circuitry is provided to make sure that both the pull uptransistors and the shorting device are not on at the same time. Themodulated backscatter transmitter “txmbs” further includes anotherantenna port “BS3” that is intended to be used when the integratedcircuit 16 is packaged in the standard SOIC package. The antenna port“BS3” provides another option for configuring a backscatter antenna. Theantenna port “BS3” supplies a one milliamp current and can drive anexternal PIN diode that would be situated between the two halves of thedipole antenna or any other suitable antenna. The other side of thatexternal PIN diode can be returned to either the antenna port “BS1” or“BS2.” Because PIN diodes are good shorting and opening devices forbackscatter applications, the transmission range of a device 12 builtwith the integrated circuit 16 can be extended over the range that isobtained using only the internal circuitry of the integrated circuit 16.This is at the expense of the need for an external component and anaccompanying increase in cost of the device 12.

FIGS. 8.07AA-BB provide a partial circuit drawing illustrating a 915 MHZtransmitter “tx915” that can be included instead of the activetransmitter described above. The transmitter “tx915” has one less stageof doubling. The chip rate also changes.

FIGS. 8.0701AA-CB provide a circuit drawing of a VCO stage modified foruse with the 915 MHZ transmitter “tx915” by adding capacitors to theoutput. The modified VCO stage is manufactured by making a metal maskadjustment employed in an alternative embodiment of the invention.

FIGS. 9AA-CB provide a circuit drawing of the analog processor“analgproc.” The analog processor “analgproc” includes a master biassource “mbs,” voltage regulators “vrg” and “vrgtx,” a bias OK circuit“biasok,” an analog port current source “aportcs,” an analog multiplexordecoder “asl,” a random number clock generator “rcg” for the pseudorandom number generator, a power up detector “pup,” and an analog todigital (A/D) converter “adanew.” The analog multiplexor decoder “asl”is an address selector used, in one embodiment, to choose from amongvarious possible inputs to the analog to digital converter. The power updetector “pup” puts out a master reset pulse upon power up. The power updetector also puts out another pulse that lasts throughout a power upcycle in which the processor performs operations appropriate upon powerup, the last operation being to reset the wake up pulse. The randomnumber clock generator “rcg” generates random numbers for use inarbitration schemes and generates a pseudo-random sequence. The masterbias source “mbs” includes a band gap regulator. The voltage regulators“vrg” and “vrgtx” generate supply voltages for various blocks ofcircuitry. The bias OK circuit “biasok” determines when the regulatorvoltage has reached a final level, and then enables the circuitry thatis driven by the regulator.

Details of Low Battery Detection

The integrated circuit 16 includes a differential I/O op-amp orcomparator comparing the voltage of the battery with a predeterminedvoltage (e.g., with band gap voltage). A low battery signal is generatedif the voltage of the battery is less than the predetermined voltage.More particularly, one of the status registers is a battery statusregister and has a value indicating if the voltage of the battery isless than the predetermined voltage. The transmitter “tx” transmits thevalue of this battery status register via radio frequency whenresponding to commands from the interrogator. In the illustratedembodiment, a battery voltage detector is shown in FIGS. 16 (and inFIGS. 11 and 9.010304AA-BB); however, the battery voltage detector canbe provided in a different location of the integrated circuit 16.

FIGS. 9.01AA-DH provide a circuit drawing of the analog to digitalconverter “ada_new.” In the illustrated embodiment, the analog todigital converter is substantially disabled and is used only to providea latch circuit for reading the low battery voltage detector. In apreferred embodiment, the analog to digital converter is used inconnection with analog sensors and to provide alarm signals whenthresholds are exceeded.

FIGS. 9.0101AA-CK provide a circuit drawing showing construction detailsof the differential I/O op-amp “dopamp” included in the analog todigital converter.

FIGS. 9.0102AA-DH provide a circuit drawing showing construction detailsof an analog divider (divide by two) “adaprescale” included in theanalog to digital converter.

FIGS. 9.0103AJ-FP provide a circuit drawing showing construction detailsof a control PLA “adactl_new” included in the analog to digitalconverter circuit of FIGS. 9.01AA-DH.

FIGS. 9.010301AA-CC provide a circuit drawing showing constructiondetails of a clock generator “adacgen_new” included in the control PLA.

FIGS. 9.010302AA-AB provide a circuit drawing showing constructiondetails of a control output driver “adacdrv_new” included in the controlPLA.

FIGS. 9.010303AA-AB provide a circuit drawing showing constructiondetails of a control output driver “adacdrvn_new” included in thecontrol PLA.

FIGS. 9.010304AA-BB provide a circuit drawing showing constructiondetails of a data latch “adadlat_new” which is included in the controlPLA and which is presently used as part of the battery voltage detector.

FIGS. 9.0104AA-DD provide a circuit drawing showing construction detailsof the analog bias circuit “adabias_new” included in the analog todigital converter.

FIGS. 9.02AA-DK provide a circuit drawing of a Vdd power up detector“pup” included in the analog processor. The power up detector puts out amaster reset pulse upon power up. The power up detector also puts outanother pulse that extends throughout a power up cycle while theprocessor performs a number of operation, the last one of which is toreset the wake up pulse. The power up detector uses a thermal voltagegenerator, which is a circuit such as is used for the low power currentcontrolled oscillator, described above. The thermal voltage generatorgenerates a small current. The power up circuit “pup” further includescurrent mirrors, and a capacitor illustrated near the top center ofFIGS. 9.02AA-DK. The current mirrors mirror the small current generatedby the thermal voltage generator. The mirrored current holds down oneside of the capacitor illustrated near the top center of FIGS.9.02AA-DK. When the power supply first rises from zero to its finalvalue, whatever that might be (e.g., 3 Volts or 5 Volts), the capacitorcouples up the bottom plate causing a signal to rise on a line “PWRUP.”The small mirrored current then slowly discharges the bottom plate until“PWRUP” switches back low. A signal from before the final inverterproducing “PWRUP” goes down to circuitry shown on the lower right ofFIGS. 9.02AA-DK. That circuitry provides a hard pull down on the bottomof the capacitor to impede any switching back and forth. After the“PWRUP” pulse, the circuitry switches the bottom of the capacitor backdown to ground much more rapidly than the small mirrored currents could.The circuitry then resets so that the only thing left holding the bottomof the capacitor low is the small current mirrored from the thermalvoltage generator. The power up detector also generates another pulse ona line “WAKEUP” shown on the right of FIGS. 9.02AA-DK. This pulse goeshigh at the same time as the pulse on line “PWRUP” but does not comedown at the same time. Instead, the pulse on the line “WAKEUP” does notcome down until the processor issues a wake up acknowledge signal online “WUACK.” The processor does not issue the wake up acknowledgesignal until completion of running of a wake up program stored in theROM.

FIGS. 9.03AA-BB provide a circuit drawing of a master bias source “mbs”included in the analog processor. The master bias source “mbs” includesa band gap reference generator “mbs_bgr” to generate bias voltages forvarious circuits of the integrated circuit 16. The master bias generatorincludes a temperature compensated current generator “mbs_cur” that isemployed in one embodiment of the invention, but is disconnected in theillustrated embodiment. The master bias source further includes areference current generator “mbs_iref” that comprises current mirrors toreplicate a reference current (e.g., 2.5 microAmps).

FIGS. 9.0301AA-DJ provide a circuit drawing showing construction detailsof a band gap reference generator “mbs_bgr” included in the master biassource. Band gap reference generators produce a reference voltage, andare known in the art. See, for example, Analysis and Design of AnalogIntegrated Circuits, Paul R. Gray and Robert G. Meyer, John Wiley &Sons. The reference voltage produced is approximately equal to the bandgap voltage of silicon, which is approximately 1.2 Volts. A band gapreference generator generates a voltage output that is independent ofpower supply and temperature.

FIGS. 9.0302AA-DI provide a circuit drawing showing construction detailsof a temperature compensated current generator “mbs_cur” included in themaster bias source.

FIGS. 9.0303AA-CF provide a circuit drawing of the reference currentgenerator “mbs_iref” included in the master bias source. The referencecurrent generator “mbs_iref” biases various circuits of the integratedcircuit 16. The reference current generator “mbs_iref” includes currentmirrors that replicate incoming current so that the reference currentgenerator can supply the same value current to a number of differentcircuit blocks.

FIGS. 9.04AA-CE provide a circuit drawing of the voltage regulator “vrg”included in the analog processor. The voltage regulator includes anop-amp having an input receiving a reference voltage “VREF” (which isapproximately 1.2 Volts). The voltage regulator further includes a largep-channel device driven by the output of the op-amp. In the illustratedembodiment, the p-channel device is made up of a plurality of p-channeldevices connected together in parallel. The voltage regulator furtherincludes an output node “VREG” driven by the plurality of p-channeldevices. The voltage regulator further includes a resistor divider,shown along the middle of the right side of FIGS. 9.04AA-CE, connectedto the output “VREG.” The resistor divider includes a fifty percentpoint (having a voltage of half of the voltage at the output node“VREG”) that is fed back to another input of the op-amp so that thevoltage at the output node “VREG” is required to be two times the inputvoltage “VREF” to complete the feedback. In the illustrated embodiment,a number of individual regulators are employed in order to isolate powersupplies to different areas of the circuit. However, in alternativeembodiments, a reduced number of voltage regulators are employed. FIGS.9.05AA-FE provide a circuit drawing of the voltage regulator “vrgtx”included in the analog processor. The voltage regulators “vrg” and“vrgtx” generate supply voltages approximately equal to two times bandgap voltage (about 2.4 Volts) for various blocks of circuitry. Thevoltage regulator “vrgtx” provides substantially the same output voltageas the voltage regulator “vrg”; however, it has a bigger drivecapability. The voltage regulator “vrgtx” is connected to the activetransmitter which requires a lot of current.

FIGS. 9.0501AA-CD provide a circuit drawing showing construction detailsof an operational amplifier without compensation “opampnc” included inthe voltage regulator.

FIGS. 9.06AA-DD provide a circuit drawing of a bias OK detector “biasok”included in the analog processor. The bias OK detector puts |out asignal indicating that regulator voltage going to the receiver is at ornear full level. The bias OK detector includes a voltage detector. Adelay is built in so that adequate time is allowed. The bias OK detectorallows biases to stabilize before releasing the clock recovery circuitand the wake up test logic.

FIGS. 9.07AA-EG provide a circuit drawing showing construction detailsof an analog port current source “aportcs” included in the analogprocessor. The analog port current source “aportcs” provides a currentwhich can be used to bias sensors external to the integrated circuit 16.The value of the current supplied by the analog port current source“aportcs” is selected from several available values by a radio frequencycommand. In the illustrated embodiment, the analog port is not employed.However, in alternative embodiments, an analog port is used.

FIGS. 9.08AA-CC provide a circuit drawing showing construction detailsof an analog multiplexer decoder “asl” included in the analog processor.The analog multiplexer decoder “asl” is an address selector. Moreparticularly, in a preferred embodiment, the analog multiplexer decoder“asl” is used to choose from among various possible analog inputs to theanalog to digital converter.

The random number clock generator “rcg” for the pseudo random numbergenerator is shown in greater detail in FIGS. 9.09AA-BB. The randomclock generator generates random numbers for use in the arbitrationscheme of the protocol to sort between multiple responding devices 12.

Details of Low Power Pseudo Random Number Generator

The device includes a random clock generator “rcg” including a linearfeedback shift register “rcg_osc” that has a plurality of stages andthat generates a pseudo-random sequence. The random clock generator“rcg” includes an oscillator “rcg_osc” that supplies clock signals tothe linear feedback shift register. The device includes a low currentgenerator, such as a thermal voltage generator, to drive the oscillatorthat supplies clock signals to the linear feedback shift register. Theshift register has two modes of operation; namely, a low power mode, anda high power mode. The random clock generator includes current mirrorsreferenced to the low current generator. In the low power mode, thecurrent to each stage of the shift register is limited by the currentmirrors. In the high power mode, the current mirror device gates aredriven to full supply voltages. This allows the shift register tooperate at a higher frequency appropriate for shifting the random numberinto the processor.

This technique is illustrated, with reference to an inverter, in asimplified schematic in FIG. 37. FIG. 37 shows a circuit includingseries connected p-type transistors 210 and 212, and series connectedn-type transistors 214 and 216 which are connected in series with thep-type transistors 210 and 212. The transistors 210, 212, 214, and 216are connected between a positive voltage “V+” and ground. The transistor210 has a gate connected to a voltage “V BIAS P” and the transistor 216has a gate connected to a voltage “V BIAS N.”

When “V BIAS N” and “V BIAS P” are controlled by a low current currentmirror (low power mode), the turn-on voltages of transistors 210 and 216are small and current through inverter transistors 212 and 214 islimited. When “V BIAS N” is pulled to “V+” and “V BIAS P” is pulled toground, the inverter operates at full speed.

FIGS. 9.09AA-BB provide a circuit drawing showing construction detailsof the random clock generator “rcg” included in the analog processor.The random clock generator “rcg” includes a low power oscillator andbias generator “rcg_osc.” The random clock generator “rcg” furtherincludes the linear feedback shift register “rcg_sreg.” The random clockgenerator “rcg” further includes a clock generator “rcg_clkgen” whichgenerates non-overlapping versions of the clock which drives the linearfeedback shift register “rcg_osc.” The linear feedback shift register“rcg_osc” generates the pseudo-random sequence. The random clockgenerator further includes circuitry (shown below the linear feedbackshift register in FIGS. 9.09AA-BB) for switching between clock schemes.This circuitry includes an n-channel device and a p-channel device(shown as circles with x's through them in FIGS. 9.09AA-BB) allowingconnection or blocking connection between the input and the output ofthe devices. The alternate clock sources are state one “S1,” state three“S3,” and phase two “P2” from the processor clock. The is shift registeris operable in a high power mode and in a very low power mode. When theprocessor wants a random number from the shift register “rcg_osc,” theseclocks are used and the shift register is operated in the high powermode to shift eight bits at a time in serial fashion into the processor.A total of sixteen bits are transferred, so two transfers of eight bitseach take place. At other times, the shift register is in the very lowpower mode and is driven by the clock generated by the low poweroscillator. In this manner, the shift register “rcg_osc” sequencesthrough its pseudo-random sequence continuously in the background untilthe shift register is called upon to provide a number.

FIGS. 9.0901AA-CH provide a circuit drawing showing construction detailsof the linear feedback shift register “rcg_sreg” included in the randomclock generator. In the illustrated embodiment, the linear feedbackshift register “rcg_sreg” is a [17,3] shift register having an output inregister seventeen. The input to the first register is the exclusive-orof registers seventeen and three. The linear feedback shift register“rcg_sreg” includes seventeen stages, so it produces a sequence of2¹⁷−1. Therefore, the odds of two devices 12 being at the same place inthe sequence are low.

FIGS. 9.090101AA-CC provide a circuit drawing showing constructiondetails of a shift register zero bit “rcg_sregbit0” included in thelinear feedback shift register. This bit is different from others sothat it can is power up in a particular state. The shift register willfunction to deliver a sequence of pseudo-random numbers as long as allregisters are not allowed to go to zero. Therefore, the zero bit“rcg_sregbit0” of the shift register is altered to guarantee that itwill be a one on power up. The shift register bit “rcg_sregbit0” shownin FIGS. 9.090101AA-CC also includes a series of n-channel and p-channeldevices to limit current in the logic gates. When the random clockgenerator is in the low power mode, the bias voltages on these seriesdevices allow only very small currents; however, when the random clockgenerator is operating in the high power mode (when the processor isshifting in a random number) then these nodes are driven to full supply.A line “BIASN” will be driven to Vdd, and a line “BIASP” will be drivento ground. Then the logic of the random clock generator operates in anormal mode.

FIGS. 9.090102AA-BB provide a circuit drawing showing constructiondetails of a shift register bit “rcg_sregbit” included in the linearfeedback shift register.

FIGS. 9.0902AA-FL provide a circuit drawing showing construction detailsof the low power oscillator and bias generator “rcg_osc” included in therandom clock generator. The low power oscillator includes a thermalgenerator, as in previously described circuitry. The low poweroscillator and bias generator “rcg_osc” further includes bias voltagegenerators shown in the middle and at the bottom in FIGS. 9.0902AA-FL.The bias voltage generator shown at the bottom in FIGS. 9.0902AA-FLincludes extra transistors to allow switching between high and low powerstates.

FIGS. 9.0903AA-CC provide a circuit drawing showing construction detailsof a clock generator “rcg_clkgen” included in the random clockgenerator.

The PN processor “pnproc” shown in FIGS. 6AA-EK is the spread spectrumprocessing circuit 40 shown in FIG. 5. The PN processor “pnproc”performs spread spectrum processing. Spread spectrum modulation isdescribed elsewhere. The PN processor “pnproc” is shown in greaterdetail in FIGS. 10AA-DD.

The PN processor “pnproc” shown in FIGS. 10AA-DD includes a digital PNcorrelator “dcorr.” The correlator receives a data stream on line“RXCHIPS” that comes from the receiver. The correlator has a thirty-onechip register and performs a comparison of the chip pattern of theincoming data stream with the expected thirty-one chip pattern. Whenthere is a total or near match, the correlator “dcorr” puts out a highsignal (a one) on line “RXDATA.” When there is a nearly total mismatch,the correlator “dcorr” puts out a low signal (a zero) on line “RXDATA.”Every thirty-one chips, “RXDATA” either changes state or does not changestate, depending on whether the PN sequence was inverted or not inverted(i.e., depending on whether a zero or one was defined by the thirty-onechip sequence). The output of the correlator “dcorr” on line “RXDATA” isa sequence of true, non-encoded, data bits of ones and zeros.

The PN processor further includes a PN lock detector “pnlockdet.” Thelock detector is a circuit that determines whether a preamble ispresent. In the illustrated embodiment, the preamble is all zeros.Therefore, the lock detector “pnlockdet” determines whether or not acertain length of zeros have occurred in a row. In the illustratedembodiment, the lock detector “pnlockdet” determines whether or not fourzeros occurred in a row. The lock detector “pnlockdet” has an outputthat is connected to the serial input output circuit “sio” in theprocessor, and enables the processor to look for the Barker or startcode.

The PN processor further includes a clock “pngclk.” The clock “pngclk”is a clock generator that is based on a clock signal “CHIPCLK” producedby the digital clock and data recovery circuit “dcr.” The clock “pngclk”puts out non-overlapping true and compliment versions of the clock andthese are used to drive circuitry in the PN processor.

The PN processor further includes a shift register “pngshr.” The shiftregister is a block of logic that can be used to generate a thirtyonechip sequence, a sixty-three chip sequence, and a two hundred andfifty-five chip sequence. A thirty-one chip sequence is always used forreceiving, but for transmitting multiple selections are available. Inthe illustrated embodiment, the integrated circuit 16 is wired to allowa selection between thirty-one and sixty-three chips. In alternativeembodiments, it can be wired to allow a selection between thirty-one andtwo hundred and fifty-five chips, or between thirty-one and sixty-threechips. In the receive mode, the PN sequence is not used explicitly,except that the middle chip and the last chip in the sequence aredetected, and those signals are used by circuitry labelled “Bit RateClock Generator” in FIGS. 10AA-DI to generate a bit rate clock for thetransmitter and receiver. Thus, the output of this shift register“pngshr” is used to generate a bit rate clock, on line “TRCLK.” In thetransmit mode, if a modulation scheme has been selected that uses spreadspectrum encoding, the output of this shift register is used to encodethe data.

The PN processor further includes a differential and PN encoder “dpenc.”The differential encoder performs differential encoding and PN encoding.The differential encoder includes an input connected to a line “TXDATA.”The data on line “TXDATA” is differential encoded by the differentialencoder, if differential encoding is selected. Both polarities ofdifferential encoding are provided for and are selected depending on thedesired modulation scheme. The differential and PN encoder can alsoimpress the PN code on the data “TXDATA” if this is selected.

The PN processor further includes a PSK/FSK generator “fskgen.” In theillustrated embodiment, PSK (phase shift keying) is performed by thePSK/FSK generator “fskgen.” In an alternative embodiment, FSK (frequencyshift keying) is performed by the PSK/FSK generator “fskgen.” Thegenerator “fskgen” has both a last chip complement output “FSKLASTCHIP”and a mid chip complement output “FSKMIDCHIP.” These outputs areconnected to the bit rate clock generator and override the outputs fromthe PN generator shift register “pngshr”. The bit rate clock generatorthen generates the appropriate bit rate clock.

The PN processor further includes D type flip-flops “pnddff,” one ofwhich is included in the bit rate clock generator.

The PN processor further includes circuitry shown on the lower right inFIGS. 10AA-DD that provides for test modes. This circuitry provides away to bring a modulating signal for the transmitter out to a digitalpad “DIGTXOUT” depending on whether an enable signal is placed on anenable pin “DIGTX.” The enable signal on enable pin “DIGTX” is alsoused, in connection with a signal on line “ForceRXON” to force thereceiver to receive in a continuous fashion. The output of the receiveris routed to a line “TESTRXDATA” and that signal is routed to a digitaloutput pad (the digital pad “DIGTXOUT” in the illustrated embodiment).

FIGS. 10.01AA-DJ provide a circuit drawing showing construction detailsof the digital PN correlator “dcorr” included in the PN processor. Thecorrelator “dcorr” includes a bias generator “dcor_bias” that generatesbias currents for other circuitry included in the correlator. Thecorrelator “dcorr” further includes a shift register “dcorr_sreg.” Theshift register “dcorr_sreg” performs a chip by chip comparison betweenthe incoming data stream and the expected thirty-one chip PN sequence.For each chip that agrees, the shift register “dcorr sreg” puts out acurrent on a line “Jagree.” For each chip that is in disagreement, theshift register “dcorr_sreg” puts out a current on a line “Idisagree.”Currents are added for each of the thirty-one chips on these lines“Iagree” and “Idisagree.” The PN correlator “dcorr” further includes twocomparator structures shown in the middle of FIGS. 10.01AA-DJ as anupper comparator and a lower comparator. The upper comparator hascurrent biasing defining a threshold, and the lower comparator hascurrent biasing defining a threshold. When a sufficient number ofcurrents flow from the shift register “dcorr_sreg” into the “Iagree”line to overcome the threshold set by the current biasing in the uppercomparator, a one is detected, and the circuit puts out a digital one.If, on the other hand, the currents in the “Idisagree” line are highenough to overcome the threshold set by the current biasing in the lowercomparator, a zero is detected, and the circuit puts out a digital zero.In other cases, the output does not change. The correlator furtherincludes circuitry shown on the right of FIGS. 10.01AA-DJ thatsynchronizes the data stream out of the correlator and into otherinformation processing circuitry.

FIGS. 10.01OIAA-BG provide a circuit drawing showing constructiondetails of the PN correlator shift register “dcorr sreg” included in thePN correlator. The shift register “dcorr sreg” performs a chip by chipcomparison between the incoming data stream and the expected thirty-onechip PN sequence. For each chip that agrees, the shift register “dcorrsreg” puts out a current on a line “Iagree.” For each chip that is indisagreement, the shift register “dcorr sreg” puts out a current on aline “Idisagree.”

FIG. 10.010101 provides a circuit drawing showing construction detailsof a PN correlator bit “dcorr_bit” included in the PN correlator shiftregister.

FIG. 10.01010101 provides a circuit drawing showing construction detailsof a shift register cell “dcorr_sregbit” included in the PN correlatorbit.

FIGS. 10.0102AA-CN provide a circuit drawing showing constructiondetails of a correlator bias generator “dcorr_bias” included in the PNcorrelator.

FIGS. 10.02AA-BE provide a circuit drawing showing construction detailsof a PN lock detector “pnlockdet” included in the PN processor. The PNlock detector “pnlockdet” detects the preamble by counting. For example,in the illustrated embodiment, the PN lock detector “pnlockdet”determines that a preamble has been received if the lock detector countsfour consecutive zeros in a row. If the PN lock detector does notachieve the four consecutive zeros, it resets and starts counting again.

FIGS. 10.0201AA-AB provide a circuit drawing showing constructiondetails of a counter bit “lockcounterbit” included in the PN lockdetector.

FIGS. 10.03AA-AB provide a circuit drawing showing construction detailsof the PN generator clock “pngclk” included in the PN processor. The PNgenerator clock is a non-overlapping clock generator.

FIGS. 10.04AA-CE provide a circuit drawing showing construction detailsof a PN generator shift register “pngshr” included in the PN processor.The PN generator shift register has select lines so that various sizedPN sequences can be generated (e.g. thirty-one, sixty-three, or twohundred and fifty-five chip sequences). The PN generator shift registeralso includes circuitry for generating mid chip and last chip signals“MIDCHIP” and “LASTCHIP” which are used for generating the bit rateclock.

FIG. 10.0401 provides a circuit drawing showing construction details ofa PN generator shift register cell “pngsreg” included in the PNprocessor.

FIGS. 10.0402AA-CB provide a circuit drawing showing constructiondetails of a PN generator shift register summer “pngssum” included inthe PN generator shift register.

FIG. 10.05 is a circuit drawing showing construction details of a PNcontroller D type flip-flop “pnddff” included in the PN processor.

FIGS. 10.06AA-DH provide a circuit drawing showing construction detailsof differential and PN encoder “dpenc” included in the PN processor. Thedifferential and PN encoder includes circuitry shown on the left inFIGS. 10.06AA-DH which performs differential encoding. The circuitryencodes data such that zeros in the incoming data cause the output totransition from either zero to one or one to zero, and ones in theincoming data cause the output not to transition. Other forms ofdifferential encoding can be performed. For example, the circuitry canencode data such that ones in the incoming data cause the output totransition from either zero to one or one to zero, and zeros in theincoming data cause the output not to transition. A selection of one ofthese two forms of differential encoding is performed by placing a highor low signal on a selection line “DIFFSEL.” Whether or not differentialencoding takes place at all is also selectable. The differential and PNencoder further includes circuitry shown on the right in FIGS.10.06AA-DH which PN encodes the data, if spread spectrum modulation isselected.

FIGS. 10.07AA-CD provide a circuit drawing showing construction detailsof a PSKIFSK generator “fskgen” included in the PN processor. ThePSK/FSK generator “fskgen” takes as its input a clock which runs at thechip rate (9.538 MHz in the illustrated embodiment). The PSK/FSKgenerator “fskgen” generates a tone for phase shift keying (e.g., 596kHz in the illustrated embodiment). The PSK/FSK generator “fskgen”further includes circuitry shown at the bottom in FIGS. 10.07AA-CD whichswitches phase according to the input data. In other words, thiscircuitry compliments ones to zeros, and zeros to ones according toinput data. If PSK or FSK is not selected, data passes through thePSK/FSK generator unaltered.

FIGS. 10.0701AA-AB provide a circuit drawing showing constructiondetails of a FSK counter bit “fskcbit” included in the PSK/FSKgenerator.

FIGS. 11AA-AB provide a circuit drawing of a battery I/O buffer “batalg”included in the integrated circuit 16. In one embodiment, batteryvoltage is compared to band gap voltage (produced by the band gapreference generator) using an op-amp. In one embodiment, the battery I/Obuffer “batalg” is used to connect a voltage to the analog to digitalconverter; however, in the illustrated embodiment, this function isperformed by a circuit “tsn.” The circuit “tsn” includes an enable line,and includes a resistor divider. When an enable signal is placed on theenable line, the resistor divider is tapped, and the output of theresistor divider goes to an op-amp for comparison with band gap voltage.

In order to detect a low battery voltage, circuitry is provided whichdefines what is a low voltage. The lowest possible value at which anindication is given that the battery voltage is low is the value atwhich the integrated circuit 16 begins to fail to operate properly.However, in a preferred embodiment, an extra margin is provided so thatthere is time to replace the battery or replace the device before theintegrated circuit 16 fails. For example, in one embodiment, the marginis 0.1 Volts. The circuitry “tsn” is therefore set up with a voltagedivider having a tap compared to the band gap voltage. The voltagedivider has resistor values selected so that when battery voltage is atthe margin (e.g. 0.1 Volts) above the lowest possible value, the tap inthe voltage divider has a voltage slightly below the band gap voltage(e.g., 1.2 Volts).

FIGS. 12AA-AB provide a circuit drawing of a digital I/O pad buffer“paddig” included in the integrated circuit 16. The digital I/O padbuffer is both an input and output buffer. The I/O pad buffer “paddig”has an input “DPAD” which is connected to a bond pad of the integratedcircuit 16. Data entering the pad buffer “paddig” from the input “DPAD”passes through an ESD protection device “esd1” and then passes on towhatever circuit for which it is an input (there are many such padbuffers “paddig” in the illustrated embodiment). Data to be output viathe pad buffer “paddig” comes into the pad buffer “paddig” via a line“DOUT” along with an enable on line “ENABLE.” The pad buffer “paddig”includes a static pull down device shown on the far right in FIGS.12AA-AB. The pad buffer “paddig” further includes n-channel andp-channel transistors shown in the right in FIGS. 12AA-AB proximate thestatic pulldown device. If an enable signal is present on line “ENABLE”and “DOUT” is high, the two p-channel devices will turn on and pull theoutput pad “DPAD” high. If “DOUT” is low, the two n-channel devices willturn on and pull the output pad “DPAD” low. The pad buffer “paddig”further includes circuitry providing for gradual pulling high or pullinglow to reduce transient currents. This is because a user may connect thepad to drive a heavy load.

FIG. 13 provides a circuit drawing of a digital input pad buffer“padigin” included in the integrated circuit 16. FIG. 13 shows the inputportion only of the pad buffer “paddig.”

FIG. 14 provides a circuit drawing of an analog I/O pad buffer “padalg”included in the integrated circuit 16. In one embodiment, the analog I/Opad buffer is used to connect an external sensor to the analog todigital converter.

Details of RF Selectable Return Link

The return link configuration logic “riconfig” provides for usercustomization of operation of the transmitter “tx.” Variouscustomizations are possible. For example, the transmitter “tx” isselectable as operating in a backscatter transmit mode, or an activetransmit mode in response to a command from the interrogator 26. This isshown in FIGS. 21 and 22. FIG. 21 is a simplified circuit schematicillustrating a transmitter “tx” switchable between an active mode and abackscatter mode, and employing separate antennas As1 and As2 for theactive mode and the backscatter mode, respectively. If the active modeis selected, the micro controller connects the antenna As1 to transmitthe output of the transmitter, using switch S1. If the backscatter modeis selected, the micro controller 34 connects the antenna As2 totransmit the output of the transmitter, using switch S2. In analternative embodiment, shown in FIG. 22, the transmitter “tx” is stillswitchable between an active mode and a backscatter mode, but employsthe same antenna 46 for both the active mode and the backscatter mode.

If the backscatter mode is selected, the interrogator 26 sends acontinuous unmodulated RF signal while the transmitter “tx” transmits aresponse to a command from the interrogator 26. The clock recovered fromthe incoming message is used to derive a subcarrier for the transmitter“tx.” In the illustrated embodiment, the subcarrier for the transmitter“tx” is a square wave subcarrier. The response to the interrogator ismodulated onto the square wave subcarrier by the device 12 using aselected modulation scheme. For example, the response can be modulatedonto the subcarrier using Frequency Shift Keying (FSK), or Binary PhaseShift Keying (BPSK).

If the active transmit mode is selected, the transmitter 32 isselectable as using amplitude modulation, or bi-phase (Binary PhaseShift Keying) modulation. The transmitter 32 is selectable as usingdifferential coding, and/or spread spectrum coding. There are variouscombinations of options that can be selected through the commands thatare sent to the integrated circuit 16 by the interrogator 26. Thetransmitter 32 is selectable as using the thirty-one chip spreadspectrum sequence, or a narrow band.

These options provide for a wide range of possible applications or usesfor the integrated circuit 16, and provide for the possibility of usingdifferent schemes in an application for different purposes. For example,an active transmit can be selected for certain purposes, while abackscatter transmit can be selected for different purposes.

FIGS. 15AA-BC provide a circuit drawing of return link configurationcontrol logic “rlconfig.” The return link configuration control logic“rlconfig” has inputs “TXSEL0,” “TXSEL1,” and “TXSEL2.” The values onthese inputs are defined by a radio frequency command sent by theinterrogator. These inputs “TXSEL0,” “TXSEL1,” and “TXSEL2” areconnected to the outputs of an output register “oreg” included in theprocessor. The return link configuration logic takes each possiblecombination of inputs “TXSEL0,” “TXSEL1,” and “TXSEL2” (there are atotal of 2×2×2=8 possible combinations) and asserts appropriate controlsignals to enable the desired return link configuration. The signalsbeing controlled by the return link configuration control logic“rlconfig” are: “ENDIL” for enabling the data interleaver; “PNOFF” forselecting whether or not PN encoding is employed for data transmitted bythe device 12; “DIFFSEL” for selecting which polarity of differentialencoding is used for transmitted data; “DIFFON” for selecting whether ornot differential encoding is employed for transmitted data; and “ENFSK”for selecting FSK (or PSK in an alternative embodiment) for transmitteddata; “BSCAT” for enabling backscatter for transmitted data; and“ENABLEAM” enables amplitude modulation.

The integrated circuit 16 further includes a number of sensors, such assensors “batalg,” “tsn,” and “mag,” in the embodiments where an A/Dconverter is included in the analog processor “anlgproc.” The sensor“batalg” is a battery voltage detector, the sensor “tsn” is atemperature sensor, and the sensor “mag” is a magnetic sensor. Thesesensors will be connected to the A/D converter in the analog processor“anlgproc” in one embodiment of the invention. In one embodiment, one ormore of these sensors are not included or not used.

Using such sensors, the device 12 can monitor things such as its ownbattery voltage, its temperature and detect the presence of a magneticfield. There are various possible uses for information sensed by suchsensors. For example, events can be counted so that, depending on theuser's application, the user can determine whether or how many times acertain item was exposed to temperature above or below a certain value(e.g., to determine likelihood of spoilage or damage). Alternatively,the user can determine whether or how many times a certain item wasexposed to a magnetic field of a certain value (e.g., when passing acertain location).

FIGS. 16AA-EH provide a circuit drawing showing construction details ofthe temperature sensor “tsn.” The temperature sensor “tsn” was designedto put out a voltage that is linearly proportional to temperature. Inthe illustrated embodiment, the circuit “tsn” has been reconfigured foruse as a low battery voltage detector.

FIGS. 16.01AA-DI provide a circuit drawing showing construction detailsof an operational amplifier “opamp” included in the temperature sensor“tsn.”

FIGS. 17AA-BB provide a circuit drawing of a magnetic field sensor“mag.” The magnetic field sensor senses magnetic fields.

FIGS. 18AA-AB provide a circuit drawing showing a chip bypass capacitor“bypcap3.” The capacitor “bypcap3” is a integrated circuit decouplingcapacitor between Vdd and ground.

FIG. 19AA-EK provide a circuit drawing of a semiconductor integratedcircuit in accordance with an alternative embodiment of the invention.The integrated circuit of FIGS. 19AA-EK is similar to the integratedcircuit shown in FIGS. 6AA-EK, like components having like componentnames, except that the integrated circuit of FIGS. 19AA-EK has no ROM,and is intended to be connected to an external ROM. This is useful fortest purposes.

FIGS. 20AA-DF provide a circuit drawing of a data processor“dataproc_t3” to be used in the integrated circuit of FIG. 19 in placeof the data processor “dataproc.” The data processor “dataproc_t3” hasan interface to external ROM.

FIGS. 20.01AA-CB provide a circuit drawing of an interface “extrom” toan external ROM.

FIGS. 20.0101AA-BB provide a circuit drawing of external ROM controllogic “extromct1” included in the interface “extrom.”

FIG. 20.0102 is a circuit drawing of an external ROM address interface“extromad” included in the interface “extrom.”

FIGS. 20.0103AA-AC provide a circuit drawing of a digital I/O pad buffer“paddigt3” included in the interface “extrom.” The digital I/O padbuffer “paddigt3” is the pad driver for the external ROM.

FIG. 20.0104 is a circuit drawing of an external ROM databus interface“extromdb” included in the interface “extrom.”

FIGS. 6AA-EK also illustrate bonding pads “PAD AA,” and “PAD A,” “PADB,” “PAD C,” “PAD D,” “PAD E,” “PAD F,” “PAD G,” “PAD H,” “PAD I,” “PADJ,” “PAD K,” “PAD L,” “PAD M,” “PAD N,” “PAD O,” “PAD P,” “PAD Q,” “PADR,” “PAD S,” “PAD T,” “PAD U,” “PAD V,” “PAD W,” “PAD X,” “PAD Y,” and“PAD Z,” which are provided around the edge of the die of integratedcircuit 16. In the illustrated embodiment, the integrated circuit 16includes a standard 20 lead SOIC package; however, any appropriateintegrated circuit package can be employed.

Connections to these pads are brought out of the package and areaccessible to the user. In this way, the user can somewhat tailor thefunction of the integrated circuit 16 to their application. In oneembodiment, however, the entire device 12 is encapsulated in a housingsuch as that shown in FIG. 3.

The pads P and Q are digital port data and clock pads, and work togetherto provide a serial input or output, or a digital connection outside theintegrated circuit. For example, if desired, data can be transmitted tothe integrated circuit 16 via radio frequency, and a response can be putout on the digital port data pad, or vice versa.

The pad R is a chip enable pad, and prevents wake up to look for anincoming radio frequency signal. There are some applications or useswhere the user knows that there will be certain periods of time when novalid radio frequency signals will be presented to the integratedcircuit 16. The user will want to prevent the integrated circuit 16 fromleaving the sleep mode so that power can be saved, and the life of thebattery 18 can be extended.

The pad S is a test mode pad for testing. When the integrated circuit 16is powered on (i.e., when power is first applied), if that pad is heldhigh then the micro controller 34 goes into a self-test mode. After theself-test, if the pad S is no longer held high, the integrated circuit16 goes to the sleep mode, and periodically awakens to look for validradio frequency signals, as it normally would. This pad S is useful tothe manufacturer of the integrated circuit 16, such as for testing priorto packaging the die of the integrated circuit 16 in the housing of theintegrated circuit 16.

The pad T is a digital transmit pad, and the pad U is a digital transmitdata pad. These pads are useful for testing. They allow the integratedcircuit 16 to operate in its intended manner, except that, if the pad Tis held high, data from the integrated circuit 16 is brought out as adigital signal on the pad U instead of being transmitted via radiofrequency using transmitter “tx.”

If the pad I is held high, data to the integrated circuit 16 is broughtin as a digital signal on the pad H instead of being received via radiofrequency using receiver “rx”. Details of the logic associated with thisfunction are included in the FIG. 8.01 in connection with lines “DIGRX”(associated with pad I) and “DIGRXDATA” (associated with pad H). Thislogic includes the NAND gates and invertors shown leading from the lines“DIGRX” and “DIGRXDATA” to a line “DataIn.”

These pads T, U, I, and H provide for testing of most functions of theintegrated circuit 16 without the need to use high frequency radiosignals. High frequency radio signals may not always be convenient in atesting lab. The pads T and U do not provide for testing of somefunctions relating to radio frequency transmission, and the pads I and Hdo not provide for testing of some functions relating to radio frequencyreception (e.g., operation of the Schottky detector). These pads T, U,V, and H do provide for testing of the spread spectrum processingcircuit 40, and for processing of protocol commands described in theappended microfiche. This allows everything but operation of radiofrequency transmitter 32 and receiver 30 to be checked prior toproceeding with that radio frequency testing. It also provides afunction for the user, in that the integrated circuit 16 does notnecessarily need to be used as a radio frequency identification device.The integrated circuit 16 has a receiver, and a transmitter, and it canbe used for various purposes, such as an actuator or beacon. If it isnot necessary to have a radio transmission or reception of data, eitherone or both form of data can be passed directly through the pads indigital form.

Note that there are separate enables T and I associated withtransmitting or receiving digital data. For example, if the digitaltransmit pad T is taken high, then the transmitter “tx” will not cause aradio frequency signal to travel to antenna 46 but instead outgoingresponses will come out on the pad U. However, the receiver “rx” willoperate normally unless the digital receive pad I is taken high.

The pad V is a TX clock pad, or transmit clock pad. Pad V was intendedto be an external input that could be used for a clock for thetransmitter 32 instead of the clock recovered from the incoming signal.In some applications, it may be necessary to have a clock that is morestable than the recovered clock, and the pad V provides a way for theuser to supply such a clock. For example, the user may connect a crystaloscillator, external to the integrated circuit 16, and that way achievea very stable carrier frequency for the transmitter “tx.” In theillustrated embodiment, pad V has been reconnected to provide a signalwhich can be used to activate an external, high performance radio.

The pads Y, Z, AA, A, and D are antenna pads for connecting the receiver30 and transmitter 32 to the shared antenna 14 or the multiple antennas44 and 46. In the preferred embodiment, circuitry that interfaces thesepads is physically located on the die next to these pads. Moreparticularly, the microwave outputs of the transmitter 32 are arrangedon the die so as to be next to (in close physical proximity to) theappropriate bond pads.

The pad B is a test RX or test receive pad, and the pad C is a test TXor test transmit pad. Because the integrated circuit 16 is usually inthe sleep mode, but wakes up briefly to look for a valid incoming radiofrequency signal, and then goes back to sleep, it can be difficult totest the receiver “rx” and the transmitter “tx.” Therefore the pads Band C provide for forcing on the receiver “rx” and transmitter “tx,”respectively, such as for testing. If a high signal is applied to thepad B, this forces the receiver “rx” to remain on. Similarly, if a highsignal is applied to the pad C, this forces the transmitter “tx” toremain on.

If the pad B is used to force the receiver on in order to exercise thecircuitry, such as through clock recovery, an input radio frequencysignal is required at the appropriate frequency (e.g., 2.45 GHz)modulated with the spread spectrum code.

The pad E is a RX input or receive input pad. This pad is connected to aside of the Schottky detector where the base band signal is available.This pad is provided for test purposes and to allow the use of ahigh-performance Schottky diode external to the integrated circuit 16.

The pad G is a VSS A pad, or analog VSS pad. The pad G is a connectionto a ground bus that only goes to the analog circuitry.

Other pads J, K, L, M, N, 0, and W are voltage supply or voltage drainpads (Vss or Vdd).

Protocol

A description of a protocol which can be employed by the device 12 forthe commands, replies, and status information is contained in a manualtitled “Micron RFID Systems Developer's Guide.” This manual relates to adevice for use with an “AMBIT” (TM) brand tracking system as well as tothe device 12. Also relevant is U.S. Pat. No. 5,500,650 to Snodgrass etal., titled “Data Communication Method Using Identification Protocol,”incorporated by reference.

Examples of commands that can be sent from the interrogator 26 to thedevice 12 are as follow:

Identify

An Identify function is used when attempting to determine theidentification of one or more of the devices 12. Each device 12 has itsown identification number TagId. It is possible that the interrogatorwill receive a garbled reply if more than one tag responds with a reply.If replies from multiple tags are received, an arbitration scheme,discussed below, is used to isolate a single device 12.

ReadAnalogPort

In one embodiment, a ReadAnalogPort function is provided which returnsthe voltage (eight-bit value) of a selected analog port on a device 12.

ReadDigitalPort

A ReadDigitalPort function returns data read from a serial port of adevice 12.

ReadTagMemory

A ReadTagMemory function returns data from a user accessible portion ofmemory included in a device 12.

ReadTagStatus

A ReadTagStatus function returns system information about a specifieddevice 12. For example, in response to this command, the device 12 willtransmit a confirmation of its TagId, a tag revision number, the lowbattery status bit, and other information.

SetAlarmMode

In one embodiment, a SetAlarmMode function is provided which is used todetermine if a set point has been exceeded on an analog port of thedevice 12 (e.g., if a sensor senses a condition exceeding apredetermined threshold). There are three alarm modes:SET_HIGH_BAND_ON_ALARM, SET_LOW_BAND_ON_ALARM, andSET_STATUS_REG_ON_ALARM.

The SET_HIGH_BAND_ON_ALARM mode sets a device 12 to a low data band, andclears a bit in the device's status register indicative of an alarmthreshold being exceeded. When a set point (threshold) is violated, thedevice 12 will switch from the low data band to a high data band.

The SET_LOW_BAND_ON ALARM mode sets a device 12 to a high data band, andclears a bit in the device's status register indicative of an alarmthreshold being exceeded. When a set point (threshold) is violated, thedevice 12 will switch from the high data band to the low data band.

The SET_STATUS_REG_ON_ALARM mode does not change data bands, but willresult in a bit ALARM_THRESHOLD_EXCEEDED in the status register beingset if the set point is violated.

SetMemoryPartition

A SetMemoryPartition function defines (initializes) a block of usermemory in a device 12 for memory partition. After being initialized, apartition may be used to store data using a function WriteTagMemory.Data may be read from the partition using a function ReadTagMemory. Thenumber of partitions available on a device 12 can be determined usingthe ReadTagStatus function.

WriteAccessId

A WriteAccessId function is used to update an access identificationAccessld for one of the memory partitions.

WriteDigitalPort

A WriteDigitalPort function is used to write data to the synchronousserial port of a device 12.

WriteTagId

A WriteTagId function is used to update the TagId of a device 12.

WriteTagMemory

A WriteTagMemory function is used to write to the user memory spaceUserMemory of a device 12.

WriteTagsRegs

A WriteTagsRegs function is used to update selected or all registers ofa device 12 including registers TagControlReg, LswTagId,TagStoredinterrId, TimedLockoutCounter, and DormantCounter for a rangeof RandomValueIds. This command can be used, for example, to disable adevice 12. If desired, the transmitter of a device 12 can be disabledwhile the receiver of that device 12 is left functional. This isaccomplished using bits KILL_TAG_0 and KILL_TAG_1 in a registerTagControlReg.

WriteTagRegsRandIdRange and WriteTagRegsTagIdRange

WriteTa RegsRandidRange and WriteTagRegsTagIdRange functions are used toupdate registers of a group of devices 12. The WriteTagRegsTagedRangefunction updates selected or all registers, including registersTagControlReg, LswTagId, TagStoredInterrid, TimedLockoutCounter, andDormantCounter, for a range of TagIds.

Examples of interrogator commands are as follows:

GetCrntAntenna

A GetCrntAntenna function returns the current antenna set used tocommunicate with a device 12.

GetCrntRetries

A GetCrntRetries function returns the number of times a command wasre-transmitted during the last tag-specific command.

GetInterrStats

A GetInterrStats function returns record-keeping parameters if theinterrogator performs this function.

GetReplyStats

A GetReplyStats function returns values that are specific to the lasttag-specific reply if the interrogator processes this information.

SetInterrRegs

A SetInterrRegs function is used to set various communication parameterson an interrogator. Not all of the parameters are used on allinterrogators.

SetInterrTest

A SetInterrTest function is used during testing. This function shouldnot be called in normal operation.

SetTimeouts

A SetTimeouts function is used to set the system watchdog timers.

A convenience command is described as follows:

IdentifyAll

An IdentifyAll function returns the number of devices 12 found withinthe system's communication range. The IdentifyAll reply parametersinclude the TagId and RandomValueId for each device 12 that isidentified.

The sequence of steps performed by a device 12 upon receipt of anIdentify command from an interrogator will now be provided, referencebeing made to FIGS. 55-57.

FIG. 55 illustrates top level steps, held in ROM, performed by the dataprocessor of the device 12 upon wake up (upon leaving a sleep mode 500)for any reason. The sleep mode is described above.

At step 502, a determination is made as to whether the device 12 is in atest mode. Test mode is enabled by holding a special pin high at powerup time. If so, the data processor proceeds to step 504; if not, thedata processor proceeds to step 506.

At step 504, a test routine is performed. The current test routinechecks the Rom, RAM, processor registers, and the timed lockout timer.After performing step 504, the data processor proceeds to step 500 (thedevice 12 returns to the sleep mode).

At step 506, a determination is made as to whether the device 12 isbeing powered up according to the status of a signal provided by a powerup detector circuit. If so, the data processor proceeds to step 508; ifnot, the data processor proceeds to step 510.

At step 508, a power up routine is performed which initializes thewakeup timer, sets up the control register, and clears the RAM. Afterperforming step 508, the data processor proceeds to step 500 (the device12 returns to the sleep mode).

At step 510, a determination is made as to whether a protocol requesthas been issued. If so, the data processor proceeds to step 512; if not,the data processor proceeds to step 514.

At step 512, the data processor executes a command processing routine.The command processing routine is described in greater detail below, inconnection with FIGS. 56A-B. After performing step 512, the dataprocessor proceeds to step 500 (the device 12 returns to the sleepmode).

At step 514, a determination is made as to whether an alarm timerrequest has been issued. This occurs once each minute. If so, the dataprocessor proceeds to step 516; if not, the data processor proceeds tostep 500 (the device 12 returns to the sleep mode).

At step 516, the data processor performs an alarm timer routine, whichin one embodiment allows a selected analog input to be compared againsta threshold. The results of the comparison can be used to set a bit andoptionally cause the chip to change data bands.

The command processing routine 512 is illustrated in greater detail inFIG. 56.

At step 518, high signals are placed on lines SIOENABLE and RFENABLE toenable the serial input output block “sio” and to enable radio frequencycommunications. After performing step 518, the data processor proceedsto step 520.

At step 520, a determination is made as to whether RFDET is highindicating that an RF signal is still present at the receiver input. Ifso, the data processor proceeds to step 522; if not, the data processorproceeds to step 524.

At step 524, the command processing routine is aborted, and the device12 returns to the sleep mode.

Steps 522, 526, 528, and 532 are used to determine whether a first byteof a command is received within a predetermined amount of time after thechip wakes up and successfully acquires the clock signal from theincoming preamble.

At step 522, a counter is initialized according to the wakeup intervalselected. After performing step 522, the data processor proceeds to step526.

At step 526, a determination is made as to whether the counter hascounted down to zero. If so, the data processor proceeds to step 524; ifnot, the processor proceeds to step 528.

At step 528, a determination is made as to whether the first byte of avalid incoming radio frequency signal has been detected. If so, theprocessor proceeds to step 530; if not, the processor proceeds to step532.

At step 532, the counter is decremented. After performing step 532, thedata processor proceeds to step 526.

At step 530, the data processor reads in a command string from theserial input output block “sio” and stores the command string in randomaccess memory. The serial input output block “sio” controls transfer ofan incoming radio frequency message from the receiver to the dataprocessor. After performing step 530, the data processor proceeds tostep 534.

At step 534, the high signals on lines RFENABLE and SIOENABLE arecleared. After performing step 534, the data processor proceeds to step536.

At step 536, the receiver is turned off in order to conserve power.After performing step 536, the data processor proceeds to step 538.

At step 538, a determination is made using CRC as to whethertransmission occurred without errors. If so, the data processor proceedsto step 540; if not, the data processor proceeds to step 524. CRC iscyclic redundancy checking, a technique known in the art used to detecterrors in transmission of data by the affirmation of error codes by boththe sending and receiving devices. In one embodiment, a check sum isused in place of a CRC.

At step 540, a determination is made as to whether the device 12 waskilled by a previous command. If so, the data processor proceeds to step542; if not, the data processor proceeds to step 544.

At step 542, a determination is made as to whether the received commandis a WriteTagRegs command which can reset the kill bits in the controlregister. If so, the data processor proceeds to step 544; if not, thedata processor proceeds to step 548, which is identical to step 524 onthe previous page of the diagram.

At step 544, a determination is made as to whether a valid command tokenexists for the received command. If so, the data processor proceeds tostep 546; if not, the data processor proceeds to step 548.

At step 546, a determination is made as to whether variables TagID andInterrID transmitted to the device 12 correctly correspond to theidentification number for the particular device 12 and theidentification number for the interrogator with which the particulardevice 12 is to correspond. If so, the data processor proceeds to step550; if not, the data processor proceeds to step 548.

At step 548, the command processing routine is aborted, and the device12 returns to the sleep mode.

At step 550, the data processor jumps to code for the specific commandthat was received by radio frequency. If the command is an Identifycommand, the data processor will jump to step 552, which is the start ofan Identify command routine.

The Identify command routine is illustrated in FIGS. 57A-B.

At step 554, a determination is made as to whether a timed lockout hasbeen set by a previously received command. If so, the data processorproceeds to step 556; if not, the data processor proceeds to step 558.

At step 556, the Identify command routine is aborted, and the device 12returns to the sleep mode.

At step 558, a determination is made as to whether a variable “InterrID”transmitted to the device 12 correctly corresponds to the identificationnumber for the interrogator with which the particular device 12 is tocorrespond. If so, the data processor proceeds to step 560; if not, thedata processor proceeds to step 556.

At step 560, Arbitration Lockout is cleared if this is requested. Afterperforming step 560, the data processor proceeds to step 562.

At step 562, a new random number is obtained if this is requested. Afterperforming step 562, the data processor proceeds to step 564.

At step 564, arbitration parameters are checked. After performing step564, the data processor proceeds to step 566.

At step 566, a determination is made as to whether the particular device12 should respond. If so, the data processor proceeds to step 568; ifnot, the data processor proceeds to step 556.

At step 568, reply parameters are assembled and stored in the RAM. Afterperforming step 568, the data processor proceeds to step 570.

At step 570, a battery status bit is updated to indicate whether thebattery voltage is below a threshold value. This information is includedin the reply to the Identify command that is sent to the interrogator.After performing step 570, the data processor proceeds to step 572.

At step 572, CRC is calculated. After performing step 572, the dataprocessor proceeds to step 574.

At step 574, high signals are set on lines RFENABLE and SIOENABLE toenable radio frequency transmission and to enable the serial inputoutput block which transfers the data to be transmitted (i.e., the replyparameters) from the processor to the transmit circuitry. Afterperforming step 574, the data processor proceeds to step 576.

At step 576, the device 12 sends a preamble, consisting of 2000 bits ofalternating pairs of ones and zeros, to the interrogator via radiofrequency. After performing step 576, the data processor proceeds tostep 578.

At step 578, the device 12 sends the 13 bit start code to theinterrogator via radio frequency. After performing step 578, the dataprocessor proceeds to step 580.

At step 580, the data processor sends a reply to the Identify command tothe interrogator via radio frequency. After performing step 580, thedata processor proceeds to step 582.

At step 582, the high signals on lines RFENABLE and SIOENABLE arecleared. After performing step 582, the data processor proceeds to step584.

At step 584, transmit mode is cleared. After performing step 584, thedata processor proceeds to step 586.

At step 586, the processor pulses the Protocol Request Acknowledgesignal which terminates the wakeup condition that initiated this entireroutine. After performing step 586, the data processor proceeds to step588.

At step 588, the data processor returns certain control register bits totheir proper states in preparation for sleep mode.

Is The processor then proceeds to step 500 and returns to sleep mode.

The sequence of steps performed by an interrogator to issue an Identifycommand will now be provided, reference being made to FIGS. 58-60.

FIG. 58 illustrates steps performed by a host processor of theinterrogator upon initialization. Initialization is started in step 600by calling a function.

At step 602, a determination is made as to whether an attempt is beingmade to open more than a maximum number of interrogators. If so, thehost processor proceeds to step 604; if not, the host processor proceedsto step 606.

At step 604, an appropriate error message is returned by setting theparameter RFID ErrorNum to the appropriate value, and a null value isreturned to the calling function.

At step 606, interrogator parameters are initialized. This includesinitializing timeout values, interrogator types and ports. Afterperforming step 606, the host processor proceeds to step 608.

At step 608, a determination is made as to whether a valid interrogatorIO port has been selected. If so, the host processor proceeds to step612; if not, the host processor proceeds to step 610.

At step 610, an appropriate error message is returned. The parameterRFID ErrorNum is set to the appropriate value and a null is returned tothe calling function.

At step 612, function addresses are assigned. This includes the functionto compute CRCs or checksums and the input and output routines. Afterperforming step 612, the host processor proceeds to step 614.

At step 614, default communication values are assigned. This includesdefault selections for diversity and communication retries. Afterperforming step 614, the host processor proceeds to step 616.

At step 616, communication hardware is reset. This initializes theinterrogator into a known state by resetting the hardware and clearingthe I/O FIFO's. After performing step 616, the host processor proceedsto step 618.

At step 618, a frequency synthesizer is initialized. This functionprograms the frequency synthesizer to the desired frequency. Afterperforming step 618, the host processor proceeds to step 620.

At step 620, a determination is made as to whether the frequencysynthesizer is programmed properly. This function is used to abort theinitialization process if the frequency synthesizer cannot beprogrammed, thereby preventing subsequent communications to occur oninappropriate frequencies. If so, the host processor proceeds to step622; if not, the host processor proceeds to step 624.

At step 622, an host memory pointer is returned that points to astructure that contains the initialized parameters. After performingstep 622, program control is returned to the Host Application Code.

At step 624, an appropriate error message is returned in the RFIDErrorNum parameter and a null is returned to the calling function.

FIG. 59 illustrates an example of a software application, starting atstep 630, that calls the Identify function and causes the interrogatorto transmit an Identify command via radio frequency.

At step 632, the function shown and described above in connection withFIG. 58 is called. After a successful call to the open functions (step632) the host computer proceeds to step 634.

At step 634, a determination is made as to whether the function shownand described in connection with FIG. 58 was successfully opened. If so,the system proceeds to step 638; if not, the system proceeds to step636.

At step 636, the host processor exits the application (or takes whateversteps are appropriate within the intended application). At step 637, theparameters are initialized for an Identify Command.

At step 638, an Identify function (described below in connection withFIG. 60) is called. After performing step 638, the host library functionproceeds to step 640.

At step 640, a determination is made as to whether a good reply wasreceived from the device 12. If so, the host computer proceeds to step642; if not, the host processor proceeds to step 644.

At step 642, reply parameters received from the device 12 are printed,displayed, or otherwise used or processed. After performing step 642,the host computer proceeds to step 646 where the application returnsresults and ends.

At step 644, the host processor exits the application or takes whateversteps are appropriate for a given application.

FIG. 60 illustrates the sequence of steps performed by the host libraryfunction at the starting at step 650, when an Identify command is issuedto the device 12.

At step 654, the command buffer is packetized, using the hostapplication initialized parameters. After performing step 654, the hostcomputer proceeds to step 656.

At step 656, the packet CRC is computed and stored at the end of thepacket.

At step 658, the packet including the CRC is stored in an interrogatortransmit queue that operates in a first in, first out fashion. Afterperforming step 658, the host computer proceeds to step 659.

At step 659, the interrogator is commanded to output the packet to theRF.

At step 660, a watchdog timer is set. After performing step 660, thehost computer proceeds to step 662.

At step 662, a determination is made as to whether a reply is availablefrom the device 12. If so, the host computer proceeds to step 668; ifnot, the host computer proceeds to step 664.

At step 664, a determination is made as to whether the watchdog timerset in step 660 has expired. If so, the host computer proceeds to step666; if not, the host computer proceeds to step 662.

At step 666, the host computer returns no reply and terminatesprocessing for the Identify command.

At step 668, CRC is checked to ensure error free transmission from thedevice 12 to the interrogator. After performing step 668, the hostcomputer proceeds to step 670.

At step 670, the reply packet is read from the reply FIFO. Afterperforming step 670, the library routine proceeds to step 672.

At step 672, the reply packet is parsed into separate parameter buffers.After performing step 672, the host library returns program control tothe host application (step 674), where processing for the Identifycommand terminates and the host application software continues.

Details of Arbitration

The arbitration of multiple interrogators per device 12 is a detectionmethod based upon each interrogator using a unique interrogator ID(InterrId). The InterrId is sent to a device 12 in a command. The device12 also stores an interrogator ID TagStoredInterrId. TheTagStoredInterrId is only updated by a WriteTagRegsXXX command. ARcvdInterrId is included in replies from a device 12. If aTagStoredInterrId does not match the RcvdInterrId then the tag will notrespond with a reply.

The arbitration of more than one tag per interrogator 26 is accomplishedby using an ArbitrationValue and an ArbitrationMask during an Identifycommand. Contained within each device 12 is a random value ID(RandomValueld) and an arbitration lockout (IDENTIFY_LOCKOUT) bit. TheRandomValueld is set to a “random” binary number upon command by aninterrogator. It may also be set by an Identify command setting aSELECT_RANDOM_VALUE bit in SubCmnd.

The following examples use a 1-byte ArbitrationValue for simplicity. Ifan interrogator 26 transmits an Identify command with itsArbitrationMask set to 0000 0000 (binary), all devices 12 in thereceiving range will respond. If there is only one device 12,communications may proceed between the interrogator 26 and device 12. Ifthere are multiple devices 12 responding, the interrogator 26 willdetect a collision and will start the arbitration sequence. To start thearbitration sequence among multiple tags, the interrogator 26 instructsthe tags to clear their IDENTIFY_LOCKOUT bit and (possibly) re-randomizetheir RandomValueld values. The ArbitrationValue 0000 0000 andArbitrationMask 0000 0001 are then transmitted to all devices 12 inrange. The devices 12 perform a logical ANDing (masking) of theArbitrationMask and the RandomValueld. If the result matches theArbitrationValue sent by the interrogator 26, the device or devices 12will reply to the Identify command. If not, the interrogator 26 willincrement the ArbitrationValue to 0000 0001 and try again.

The interrogator 26 then checks each of the possible binary numbers(0000 0000 and 0000 0001 in this case) in the expanded mask (0000 0001)for a response by a device 12. If a single device 12 responds to one ofthese values, the interrogator 26 will reply by commanding it to set itslockout bit. If any collisions are detected at this mask level, the maskwould be widened again by one bit, and so on through the eight bit widemask (256 numbers). If no collisions are detected for a particularArbitrationValue and ArbitrationMask combination, the TagId returned inthe reply is used for direct communication with that particular device12. During the arbitration sequence with up to about one hundred devices12, the mask will eventually grow large enough such that all devices 12can respond without collision. After the mask widens to four or fivebits, more devices 12 have unique random numbers and single tag repliesare received. Thus with each expansion of the ArbitrationMask, there arefewer and fewer tags left to Identify.

With a large number of tags in range, it is possible that severaldevices 12 will choose the same value for their RandomValueld. In thiscase, the complete mask will be used. Collisions will still occur andthe remaining tags will be instructed to select a new Random ValueId. Ifan application dictates, for example, that one hundred tags will usuallybe present in range of the interrogator 26, it would be advantageous tostart with the mask set to eight bit wide (11111111) and count upthrough 256 instead of starting with the mask set at 0000 0000, followedby 0000 0001, 0000 0011, etc. Other arbitration schemes can beimplemented by the user.

Applications

There are a large number of possible applications for devices such asthe device 12. Because the device 12 includes an active transponder,instead of a transponder which relies on magnetic coupling for power,the device 12 has a much greater range.

One application for devices 12 is inventory control to determine thepresence of particular items within a large lot of products.

Another application for devices 12 is electronic article surveillance(EAS). The devices 12 can be attached to retail items in a store havingan interrogator 26 at the exits, for detection of unauthorized removalof retail items. The merchant can deactivate or remove devices 12 fromretail items for which proper payment has been made.

Another application for devices 12 is to track migration of animals.

Another application for devices 12 is to counteract terrorism bymonitoring luggage entering a plane to ensure that each item of luggagethat enters the plane is owned by a passenger who actually boards theplane. The devices 12 can also be used to monitor luggage to locate lostluggage.

The device 12 can be use to track packages, such as courier packages.

The device 12 can be used to track hazardous chemicals or waste toensure that it safely reaches a proper disposal site.

The device 12 can be used for security purposes, to track personnelwithin a building. The device 12 can also be used for access control.

The device 12 can be used to monitor and manage freight transit. Forexample, interrogators 26 can be placed at the entrance and exit of aterminal (e.g., a rail or truck terminal), to monitor incoming andoutgoing shipments of vehicles bearing the devices 12.

The device 12 can be used to impede car theft. A European anti-theftdirective (74/61/EEC) provides that all new car models sold afterJanuary 1997 must be fitted with electronic immobilizers and approvedalarm systems. The devices 12 can be provided on keychains or within carkeys, and interrogators 26 placed in cars, so that the vehicle will beinoperable unless the specified device 12 for a specific car is used.The interrogator 26 can control the door locks of a car, or the ignitionof the car, or both. Because the device 12 includes memory, theinterrogator 26 in the car can periodically automatically change valuesin the device 12 (like changing a password).

Devices 12 can be placed in cars and used in connection with electronictoll collections systems. Because the devices 12 can be used to identifythe respective cars in which they are placed, interrogators 26 in tollplazas can charge appropriate accounts based on which cars have passedthe toll plaza.

Devices 12 can be placed in cars and used in connection with parkingsystems. Because the devices 12 can be used to identify the respectivecars in which they are placed, interrogators 26 in parking areas candetermine when a vehicle arrives and leaves a parking area.

The devices 12 can be used for inventory control of rental equipment.

The devices 12 can be used where bar code labels will not properly workbecause of harsh environmental conditions (e.g., grease, dirt, paint).

In compliance with the statute, the invention has been described inlanguage more or less specific as to structural and methodical features.It is to be understood, however, that the invention is not limited tothe specific features shown and described, since the means hereindisclosed comprise preferred forms of putting the invention into effect.The invention is, therefore, claimed in any of its forms ormodifications within the proper scope of the appended claimsappropriately interpreted in accordance with the doctrine ofequivalents.

What is claimed is:
 1. A frequency doubler comprising: a first cellincluding a first pair of transistors having sources that are coupledtogether, a second pair of transistors having sources that are coupledtogether, a first one of the transistors of the first pair and a firstone of the transistors of the second pair having gates coupled together,a second one of the transistors of the first pair and a second one ofthe transistors of the second pair having gates coupled together, thefirst transistor of the first pair and the second transistor of thesecond pair having drains coupled together, the second transistor of thefirst pair and the first transistor of the second pair having drainscoupled together, the first cell further including a third pairincluding first and second transistors having sources coupled together,the first transistor of the third pair having a drain coupled to thesource of the second transistor of the first pair, the second transistorof the third pair having a drain coupled to the source of the secondtransistor of the second pair, and the first cell further including acurrent source coupled to the sources of the third pair and forwardbiasing the third pair; and a second cell including a first pair oftransistors having sources that are coupled together, a second pair oftransistors having sources that are coupled together, a first one of thetransistors of the first pair of the second cell and a first one of thetransistors of the second pair of the second cell having gates coupledtogether, a second one of the transistors of the first pair of thesecond cell and a second one of the transistors of the second pair ofthe second cell having gates coupled together, the first transistor ofthe first pair of the second cell and the second transistor of thesecond pair of the second cell having drains coupled together, thesecond transistor of the first pair of the second cell and the firsttransistor of the second pair of the second cell having drains coupledtogether, the second cell further including a third pair including firstand second transistors having sources coupled together, the firsttransistor of the third pair of the second cell having a drain coupledto the source of the second transistor of the first pair of the secondcell, the second transistor of the third pair of the second cell havinga drain coupled to the source of the second transistor of the secondpair of the second cell, and the second cell further including a currentsource coupled to the sources of the third pair of the second cell andforward biasing the third pair of the second cell, the second transistorof the third pair of the second cell having a gate defining a thirdinput node of the second cell, and the first transistor of the thirdpair of the second cell having a gate defining a fourth input node ofthe second cell; and the drain of the second transistor of the firstpair of the second cell being coupled to the drain of the secondtransistor of the first pair of the first cell, the drain of the secondtransistor of the second pair of the second cell being coupled to thedrain of the second transistor of the second pair of the first cell;wherein the first cell is configured to receive a first sinusoidal wave,wherein the second cell is configured to receive a sinusoidal waveshifted from the first sinusoidal wave, and wherein no integrator isemployed in the frequency doubler.
 2. A method of doubling frequency,the method comprising: coupling first and second ceils together, thefirst cell including a first pair of transistors having sources that arecoupled together, a second pair of transistors having sources that arecoupled together, a first one of the transistors of the first pair and afirst one of the transistors of the second pair having gates coupledtogether, a second one of the transistors of the first pair and a secondone of the transistors of the second pair having gates coupled together,the first transistor of the first pair and the second transistor of thesecond pair having drains coupled together, the second transistor of thefirst pair and the first transistor of the second pair having drainscoupled together, the first cell further including a third pairincluding first and second transistors having sources coupled together,the first transistor of the third pair having a drain coupled to thesource of the second transistor of the first pair, the second transistorof the third pair having a drain coupled to the source of the secondtransistor of the second pair, and the first cell further including acurrent source coupled to the sources of the third pair and forwardbiasing the third pair; and the second cell including a first pair oftransistors having sources that are coupled together, a second pair oftransistors having sources that are coupled together, a first one of thetransistors of the first pair of the second cell and a first one of thetransistors of the second pair of the second cell having gates coupledtogether, a second one of the transistors of the first pair of thesecond cell and a second one of the transistors of the second pair ofthe second cell having gates coupled together, the first transistor ofthe first pair of the second cell and the second transistor of thesecond pair of the second cell having drains coupled together, thesecond transistor of the first pair of the second cell and the firsttransistor of the second pair of the second cell having drains coupledtogether, the second cell further including a third pair including firstand second transistors having sources coupled together, the firsttransistor of the third pair of the second cell having a drain coupledto the source of the second transistor of the first pair of the secondcell, the second transistor of the third pair of the second cell havinga drain coupled to the source of the second transistor of the secondpair of the second cell, and the second cell further including a currentsource coupled to the sources of the third pair of the second cell andforward biasing the third pair of the second cell, the second transistorof the third pair of the second cell having a gate defining a thirdinput node of the second cell, and the first transistor of the thirdpair of the second cell having a gate defining a fourth input node ofthe second cell; and the drain of the second transistor of the firstpair of the second cell being coupled to the drain of the secondtransistor of the first pair of the first cell, the drain of the secondtransistor of the second pair of the second cell being coupled to thedrain of the second transistor of the second pair of the first cell,wherein input signals are applied to the first and second cells withoutuse of an integrator.
 3. An RFID transponder including a frequencymultiplier, the frequency multiplier comprising: a first cell includinga first pair of transistors having sources that are coupled together, asecond pair of transistors having sources that are coupled together, afirst one of the transistors of the first pair and a first one of thetransistors of the second pair having gates coupled together, a secondone of the transistors of the first pair and a second one of thetransistors of the second pair having gates coupled together, the firsttransistor of the first pair and the second transistor of the secondpair having drains coupled together, the second transistor of the firstpair and the first transistor of the second pair having drains coupledtogether, the first cell further including a third pair including firstand second transistors having sources coupled together, the firsttransistor of the third pair having a drain coupled to the source of thesecond transistor of the first pair, the second transistor of the thirdpair having a drain coupled to the source of the second transistor ofthe second pair, and the first cell further including a current sourcecoupled to the sources of the third pair and forward biasing the thirdpair; and a second cell including a first pair of transistors havingsources that are coupled together, a second pair of transistors havingsources that are coupled together, a first one of the transistors of thefirst pair of the second cell and a first one of the transistors of thesecond pair of the second cell having gates coupled together, a secondone of the transistors of the first pair of the second cell and a secondone of the transistors of the second pair of the second cell havinggates coupled together, the first transistor of the first pair of thesecond cell and the second transistor of the second pair of the secondcell having drains coupled together, the second transistor of the firstpair of the second cell and the first transistor of the second pair ofthe second cell having drains coupled together, the second cell furtherincluding a third pair including first and second transistors havingsources coupled together, the first transistor of the third pair of thesecond cell having a drain coupled to the source of the secondtransistor of the first pair of the second cell, the second transistorof the third pair of the second cell having a drain coupled to thesource of the second transistor of the second pair of the second cell,and the second cell further including a current source coupled to thesources of the third pair of the second cell and forward biasing thethird pair of the second cell, the second transistor of the third pairof the second cell having a gate defining a third input node of thesecond cell, and the first transistor of the third pair of the secondcell having a gate defining a fourth input node of the second cell; andthe drain of the second transistor of the first pair of the second cellbeing coupled to the drain of the second transistor of the first pair ofthe first cell, the drain of the second transistor of the second pair ofthe second cell being coupled to the drain of the second transistor ofthe second pair of the first cell, wherein no integrator is employed inthe frequency multiplier.
 4. A frequency doubler comprising: a firstGilbert cell having first and second inputs; a second Gilbert cellhaving third and fourth inputs, coupled to the first Gilbert cell; thefirst Gilbert cell being configured to receive first and secondgenerally sinusoidal input signals at the first and second inputs,respectively, the second input signal being generally shifted 180° fromthe first input signal, the second Gilbert cell being configured toreceive third and fourth generally sinusoidal input signals at the thirdand fourth inputs, respectively, the third input signal being generallyshifted 90° from the first input signal, the fourth input signal beinggenerally shifted 270° from the first sinusoidal input, the frequencydoubler having first and second outputs and being configured to providefirst and second generally sinusoidal output signals at the first andsecond outputs, respectively in response to the first, second, third,and fourth inputs, the second output signal being generally shifted 180°from the first output signal, wherein the first and second outputsignals have generally the same average output values without need foran integrator between the first and second inputs and third and fourthinputs.